390411-1-F FINAL REPORT TO TEXAS INSTRUMENTS Attn: Dr. Shashi Sanzgiri COUPLING THROUGH SLOT APERTURES Submitted by Linda P.,B. \\Katehi Associate Professor Department of Electrical Engineering and Computer Science The University of Michigan Ann Arbor, MI 48109-2122 Period of Study: October 15, 1990-June 30, 1992 Cost: $20,000 Date of Submission: June 30, 1992.

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FINAL REPORT 1. TEXAS INSTRUMENTS Award: EECS 390411 2.Period Covered By This Report: October 15, 1991- June 30, 1992 3.Title of Proposal: Coupling Through Slot Apertures 4.Name of Institution: University of Michigan 5. Principal Investigator: Linda P.B. Katehi 7. List of Manuscripts Published or Submitted for Publication/Presentation Under Texas Instruments Sponsorship During This Reporting Period: 1. Norman L. VandenBerg, "Full-wave Analysis of Microstrip-fed Slot Antennas and Couplers," Ph.D Dissertation, The University of Michigan, Ann Arbor, October 1991. (Appendix A) 2. Norman L. VandenBerg and Linda P.B. Katehi, "Broadband Vertical Interconnects Using Slot-Coupled Shielded Microstrip Lines," IEEE Trans. on Microwave Theory and Techniques, Vol.40, No.1, January 1992, pp.81-88. (Appendix B) 3. Norman L. VandenBerg and Linda P.B. Katehi, "Generalized FullWave Treatment of Shielded Multi-Layer Structures Using the Vector Wave Function Field Expansion Method," submitted to the Electromagnetics. (Appendix C)

8. List of Interim Reports Generated From This Work: 1. Norman L.VandenBerg, "Investigation of Strip-Fed Double-Tuned Slot Coupling Through Waveguide Sections," August 6,1990. (Appendix D) 2. Norman L.VandenBerg, "Full-Wave Analysis of Aperture Coupled Shielded Microstrip Lines," October 15, 1990. (Appendix E) 3. Norman L.VandenBerg, "Proposed Dimensions for Cavity-Backed Fixture Measurements," October 24, 1990. (Appendix F) 4. Norman L.VandenBerg, "Investigation of a Stripline Slot Coupler," January 16, 1991. (Appendix G) 5. Norman L. VandenBerg, "Report of Progress Analysis of Finite Thickness Slots," January 29, 1991. (Appendix H) 6. Norman L. Vandenberg, "Corrected Results for Stripline Slot Coupler," March 7, 1992. (Appendix I)

INTRODUCTION The research conducted during this study has concentrated on the following projects: a) Study of Broadband Vertical Interconnects Using Aperture Coupled Shielded Microstrip Lines b) Thick Slots c) Full-Wave Analysis of Microstrip Fed Slot Antennas. In addition to Texas Instruments support, all of the above projects have been partially supported by NSF as part of the PYI matching funds program. The progress in each one of these projects is summarized below. STUDY OF BROADBAND VERTICAL INTERCONNECTS USING APERTURE COUPLED SHIELDED MICROSTRIP LINES Appendix A Appendix E Appendix B Appendix G Appendix C Appendix I A full-wave space-domain integral equation analysis of aperture coupled shielded microstrip lines has been performed based on the Equivalence Principle. The formulation has the capability to model multi-layered substrates through the derivation of the associated dyadic Green's function which represents the layers through impedance boundary conditions (Appendix C). The method of Moments is used to solve for the line currents and slot voltage with even and odd mode excitations which are then interpreted through transmission line analysis to determine the two-port scattering parameters. A parametric study together with experimental data is presented which demonstrates the behavior of the coupler and the accuracy of the technique (see Appendices A,B).

THICK SLOTS Appendix A Appendix D Appendix H It has been suggested that strip-fed radiating slots may play an important role in the recent trend towards more complex antenna systems. This is particularly true for monolithic phased array applications where active devices are integrated with the radiating elements for phase control and in some cases to generate power. One of the practical difficulties which arises with the introduction of active devices is the dissipation of heat. One solution is to increase the thickness of the antenna face for use on the electrical design, however, since the slots may no longer be considered infinitely thin as is commonly done. On the other hand, we can take advantage of the thickness to improve bandwidth, which has traditionally been a fundamental limitation of slot arrays. This possibility was the primary topic of the performed study. While semi-empirical and approximate correction factors for finite thickness slots have been introduced and may be adequate and appropriate in some circumstances, it is desirable to develop exact methods to improve accuracy and generality of numerical design tools. The full-wave integral equation technique which has been presented is also applicable to this class of antennas, allowing not only the analysis of thick slots but also slots which couple through a section of waveguide as shown in Appendix A, Figure 7.1, and Appendices D,H. This latter case also provides the mechanism for improved performance. In the study performed, the rectangular waveguide section connecting the slots has been adjust to provide higher overall bandwidth.

FULL-WAVE ANALYSIS OF MICROSTRIP FED SLOTS Appendix A Appendix C Appendix F Recently, complex systems have imposed new antenna requirements with demands for compact conformability, reduced weight, and higher order antenna functions such as electronic beam stearing, polarization control and power generation. A suitable alternative approach employing stripline-fed series slots isolated from higher order modes by shorting pins has been widely applied in the past. However, despite the advantages of such an arrangement wide use has been hindered by the lack of accurate computational models. The necessity for such models is intensified by the narrow-bandwidth properties of these slot radiators. During the reporting period, we have studied the properties of these slot antennas and the effect of the pin curtains on the resonant characteristics (see Appendix B) and we have investigated ways to increase their bandwidth.

APPENDIX A Norman L. VandenBerg Full-Wave Analysis of Microstrip-Fed Slot Antennas and Couplers

FULL-WAVE ANALYSIS OF MICROSTRIP-FED SLOT ANTENNAS AND COUPLERS by Norman L. VandenBerg Linda P. B. Katehi Radiation Laboratory Department of Electrical Engineering and Computer Science The University of Michigan Ann Arbor, MI 48109-2122 Report No. 877 August 13, 1991

This report is the unabridged Ph.D. thesis of Dr. Norman L. VandenBerg of tihe Department of Electrical Engineering and Computer Science, the University of MNichigan. The research was supported by the second author through a Graduate Student Research Assistantship with funding from various sources.

ACKNOWLEDGEMENTS Before presenting the material in this document whiclh is a summarx of some of the work I have performed over the past four years, I would like to thank and acknowledge the many people and organizations who have contributed to and supported tis researcli. First, I would like to express my appreciation to the Radiation Laborntory and C'entcr for Space Terahertz Technology for financial support in thle foriiil of a Researcli Assistantship. In particular, I thank Professors Val Liepa and Cllen-To Tai for support in the first year of my return to the University, and Professor Linda liKatehi for the remniainder of my stay. A special debt of gratitude is owed to Professor Tai for many years of teacling and encouragement. My first course with Professor Tai was a Junior level course on Vector Calculus in my undergraduate days and since then I hIave lhad the priviledge of attciidillg three other courses under his tutelage, including his course on Dyadic Green's Functions in Electromagnetic Theory. As the reader may note, Professor Tai's approach has had a strong influence on my work and I hope that I have been able to contribute in at least a small way to the methodology he has pioneered. I also would like to thank many of my associates who have contributed to this effort. I especially thank Dr. William Harokopus and Mr. Andrew Engel, not only for their friendship but their technical contributions as well. I also thank other members of our group, including Miss Emilie VanDeventer and Messrs. Nihad Dib and George Eleftheriades for numerous helpful discussions. In addition, I would like to express my appreciation to

M,r. Jeffrey Collins, Drs. Kamal Sarabandi, Kasra Barkeshli. Jianming Jin. Mlark Rico!. Leland Pierce anld many others too numerous to mention for their contributions throu!li both teclhnical disciissions and camaraderie. A special mention is deserved by Dr. Harry LaFuse at the Allied-Bendix Guidance Svstems Divisioln - Mishawaka, whose generous contribution of many classic text and reference books on my departure to return to graduate studies, has made an invaluable contribujtion to tlis work. Also, I am grateful to the Allied-Bendix Communications Division for their support in Iiiaking many fixtures and measurements for the verification of the slot antenna models, and to Texas Instruments for their similar support. Finally I would like to thank my committee members for their helpful suggestions and careful review of the manuscript. I especially acknowledge my advisor, Dr. Linda Katehi, for lier support and encouragement; she has been a constant source of ideas and insights whlicih lave greatly contributed to this work.

PREFACE This dissertation expounds on the thesis that numerical methods and models ca be ((developed which are sufficiently accurate to enable a microwave engineer to design microst rip)fed slot devices, without resorting to the empirical or semi-empirical techniques characteristic of more traditional methods. This possibility is of importance since the traditional approaches are typically time-consuming and therefore expensive. Tile advalntages of nilmerical models are widely recognized, not only in the microwave industry, btut ill IIllAll other fields as well. Withil thle rapid development of personal computers and workstatiolls in terms of availability and processing power, computer-aided design (CAD) capability, becoming even more important for the development of technology. Slots have been used in microwave designs for many years, particularly in waveguide- fed antenna arrays. Stripline, and more recently, microstrip Line has also been used frelquently. especially when there are active microwave components involved. It is now recognized tlhat combinations of these types of structures and devices offer many significant advantagcs and will be needed to meet the requirements for advanced, state-of-the-art systems wllichl have recently been proposed. Additional changes in technology have created the need for the ability to analyze structures with multi-layered substrates and superstrates and. increasingly, there has been a pushl toward higher and higher operating frequencies. For these reasons, we have developed thre analytical and numerical methods to be presented here. It will be shown that many of these issues can be addressed in terms of computer aided design and a considerable advancement and improvement over previous work 1in

area tias been achieved. In Cltapter I. we begi,i by introducing various historical aspects of tile use of slots in microwave antennas. Tlhrouglh an examination of these devices in tile context of present thinking, we will define the types of problems to be analyzed in this work; specificall. microstrip-fed slot antenna elenients and couplers. The basis for the numerical models wil be formulated in terms of initegral equationis. Full-wave analysis by means of exact Greeni's funictions is used with a view toward application of these methods at high frequencies where other methods generally fail. Chliapter 11 will presenit tile derivations of the necessary dyadic Green's functions which will be used throughoiit the remainder of the work. The approach will use a'field expansion method' in terms of vector wave funictions which will be explained and defined. The method of scattering superposition will be used and a methlod, not previously presented for this approach, etnploying impedance boundary conditions and field matching proceedures will be developed. The entire approach is in contrast to the more widely used, and perhaps more familiar,'vector potential method'. Some comparisons to and deficiencies with past usages of the latter approach will be pointed out. To further illustrate the differences, the method of scattering superposition with impedance boundary conditions as applied to the vector potenttial methlod will be illustrated by example in the appendix, since even for this methlod, there are significant advantages that have not previously been presented. The application of the method of moments to the integral equations is detailed in Chapter III. The treatment will be generalized to include variations on the main thrust of the work, to show low slots and lines can be modelled with arbitrary orientations relative to each other and the shielding structure. Although the applications discussed in later chapters impose some simplifying assumptions on the geometry, the material here lays the groundwork for further extensions to the work which may be implemented at a later date.

Also iiicluded inl Cliapter III is some discussion of mathematical relations and progranimtli teclniques whlich Iave been used to greatly reduce the computation time for generating the required matrix elements. Several applications are introduced in Chapter IV. This chapter is devoted to the discussion of problems which can be reduced to two dimensions. The first part demonstrates the procedures for treatment of layered structures through analytical field matching tlhroughout the layers. Applications discussed include the evaluation of transmission liiie parameters for microstrip lines and the visualization of field behavior for both shielded strip and slot lines. The accuracy of the technique is verified for microstrip by comparing to data available in the literature and a commercial computer-aided design package. The second part deals with the development of a model for scattering from vertical wires in waveguides. For small diameter wires, the model can be greatly simplified compared to approaches used in the literature. The validity of the model is verified by comparison to experimental measurements. The motivation for the work in this chapter is to support the modelling of applications discussed in Chapter VI. Chapter V presents the analysis of microstrip-fed slot couplers. Expressions for the S-parameters which characterize their behavior are derived based on network analysis of even- and odd-mode excitations of the structure. The procedure is referred to as the'Standing Wave Metlhod' and involves an interpretation of the method of moments solution for currents on the microstrip lines. This has become a sort of'standard' approach to a variety of similar problems but has some drawbacks as will be pointed out. Also presented in this chapter is a discussion of the fixture design and experimental results which verify the accuracy and validity of the method. Chapter VI introduces the radiating slot or antenna element problem by deriving an alternative approach fot finding network parameters. The method is based on an application

of the Reciprocity Theorem and is here referred to as the'Reaction Tlheorem.Netliod' or'Reaction Method'. The requirement for a new approach is a consequence of the limitationl> of the Standing Wave Method discussed in Chapter V. Historically, the Reaction Mlethod was introduced many years ago for application to simple waveguide-fed slots; however. for our use, the structures are more complex and as a result, the application of the netlhod is also. The details of how the technique is applied are discussed in tlhis cliapter. together with experimental results which demonstrate the capability of the nunierical methods. In Chapter VII, a variation of the radiating slot is explored. In this case, a slot of finite thickness is introduced. Instead of simply making the slot thicker. which can also be modeUed by this approach and has been treated through more approximate nmethods in thle past, the case where tie strip-fed slot couples to the radiating slot tllrougli anl intervellintg section of rectangular waveguide is presented. In this case, we find the important result that the slots can be detuned to extend the bandwidth of the element. The dissertation concludes with Chapter VIII which summarizes the techniques developed. The points where the effort is judged to have succeeded are outlined as well as where the numerical models fail. In the latter cases, the suspected causes for deficiencies in the approach are discussed along with recommendations for remedies. These issues form thle basis of suggestions for the extension of this work and exploration of related areas. Specifically, the treatment of a'T-bar' fed slot is discussed. This is a slot fed by a microstrip line terminated in a T-junction, whose branch arms are short-circuited to the cavity side walls. A similar device has been previously shown to provide extended bandwidthl for cavity-backed aperture antennas. A few final comments about the mechanics of this work: A number of programs were developed for the numerical modelling and analysis of the structures discussed as well as for post-processing the data. These programs were written almost exclusively in FORTRAN

and run on a variety of maclines. includiig the Univcrsity of Mtichigan IB3NM 3090,i600E mainframe and IBM RSG000 workstations for thle numerically initensive operatioiis. majority of thle remaining processing was performed on HP/Apollo workstations, primarily a DN2500. The manuscript was typeset using LATEX text processing software with macros developed at the University of Nlichigan for dissertation formats. Rectangular twodimensional plots used throughout were produced in PostScript by a menu-driven plotting program developed jointly by the author and Dr. Leland Pierce. Smith Chart plots were produced by simnilar programs developed by the author. Most of the drawings were generated using eitlier XFig or directly witlh PostScript and incorporated in PostScript form.

TABLE OF CONTENTS DEDICATION...................... ACKNOWLEDGEMENTS..m............... ii PREFACE................. LIST OF FIGURES..................... xiii LIST OF TABLES................... xvii LIST OF APPENDICES................ xviii CHAPTER I. INTRODUCTION........................... 1.1 Motivation and Background. 1.2 General Description of Analysis Approach.......... 1.3 Integral Equations and Notation.................. (i 1.3.1 Dyadic Green's Functions for Physical Quantities.. 1.3.2 Dyadic Green's Functions for'"Dual" Quantities...... s 1.3.3 Integral Solutions to Helmholtz Equations using Dyadic Green's Functions....................... 10 1.4 Formulation by Application of Boundary Conditions......... 12 II. DYADIC GREEN'S FUNCTIONS.................... 18 2.1 Impedance Boundary Conditions for Layered Structures....... 19 2.2 Dyadic Green's Functions for an Infinite Covered Half Space-.... 22 2.2.1 Magnetic Current....................... 22 2.2.2 Electric Current..................31 2.3 Dyadic Green's Functions for Layer Filled Rectangular Waveguides 33 2.3.1 The Parallel Plate Green's Function.......... 33 2.3.2 Rectangular Waveguide with Electric Currents...... 37 2.3.3 Rectangular Waveguide with Magnetic Currents.39 2.4 Dyadic Green's Functions for Layer Filled Rectangular Cavities.. 40 2.4.1 Homogeneously Filled Rectangular Waveguide: TE and TM Modes......................... 40 2.4.2 Cavity with Opposing Impedance Walls: Electric Current 43

2.4.3 TE and TM lModes in IHomogellcously Filled Rcc:an:.:! Wvaveguide and Cavities with Opposing Impedance \:Wa' Magnetic Currents..............' III. METHOD OF MOMENTS FORMULATION........... - 3.1 Definition of Coordinate Systems and Basis Functions...... 3.2 Excitation Models....... 53 3.3 Expansion of the Dyadic Green's Functions for tile Cavitsy. 3.4 Identification and Reduction of thle Integrands.... 61.. 3.5 Integration........................................... 64 3.6 Numerical Evaluation Considerations..6......... 6 3.6.1 Preconlmputatioll.............. 66 3.6.2 Transformnations................6.. 7 G 3.6.3 Convergence, Algorithms and Run-Time......... G9 IV. ANALYSIS OF TWO-DIMENSIONAL STRUCTURES...... 4.1 Application to General.Multi-layered Shielded Microstrip Structures 78 4.1.1 LSE Modes.81. 4.1.2 LSM Mllodes...................... 83 4.1.3 Cross Tcrmins.................... S 4.1.4 Integratio,n of Power and Reaction terms......... S5 4.1.5 Applications....................... 4.2 Application to Multi-layered Slotline.......... 93 4.2.1 LSE Modes........................... 93 4.2.2 LSM Modes........................... 95 4.2.3 Cross Terms............................ 96 4.2.4 Application to Finline and Suspended Finline....... 97 4.3 Scattering from Pins in Rectangular Waveguide........... 99 4.3.1 Reflection Coefficient Formula Derived from the Reciprocity Theorem............................ 100 4.3.2 Method of Moments Formulation............. 102 4.3.3 Validation........................... 103 4.4 Summary and Conclusions....................... 113 V. COUPLING THROUGH STRIP-FED SLOTS........... 114 5.1 Network Analysis...................1......... 114 5.2 Fixture Design.............................. 117 5.3 Numerical and Experimental Results......... 118 5.4 Summary................................... 126 VI. ANALYSIS OF STRIP-FED RADIATING SLOTS........ 128 6.1 Equivalent Slot Impedance....................... 130 6.2 Numerical Results and Measurements................ 132

6.3 Summary.................. VII. MODELLING OF THICK-SLOTS........... 7.1 Numerical Results for Waveguide Coupled Slots........... L14 7.2 Summary and Conclusions..................... 153 VIII. CONCLUSIONS AND RECOMMENDATIONS FOR FUTURE WORK...................................... 154 8.1 Summary of Achievements.................... 155 8.1.1 Two-Dimensional Analysis....155 8.1.2 Couplers......................... 156 8.1.3 Radiating Slots.............. 157 8.1.4 Modelling of Thick Slots.............. 158 8.2 Model Limitations.......................[.. 159 8.3 Recommendation for Future W'ork............... 160 APPENDICES................................164 BIBLIOGRAPHY.............................. 191 xii

LIST OF FIGURES Figure 1.1 A hypothetical integrated antenna employing nuimerous slot couplinIg str ictures................. 4 1.2 Cavity layout and equivalent magnetic curreiits for a radiating slot... 13 1.3 Electric and equlivalent magnetic currents for nmicrost rip-to- nicrosttrip coupler with a'thick' slot........................ 17 2.1 Impedance boundary condition represeiitation of a nmll i-layered structure.'21 2.2 Layered infinite space as represented by impedance boundary conditions. 26 2.3 Parallel plate waveguide coordinate system.................3. 34 3.1 Cavity (, y) and Strip-fixed (v,v) Coordinate System........... 48 3.2 Current Expansion Functions....................... 49 3.3 Slot-fixed Coordinate System ((, ().................... 50 3.4 Convergence behavior for several parameters of a centered sliielded microstrip transmission line. Dimensions a = b =.25, w = h =.025, E, = 9.7. Terminal values at 1000 modes were Zo = 49.63Q, f,.eff. = 6.90, Hx = 12.4.5A/A,o............................ 70 3.5 Convergence behavior for peak normalized resistance versus maximum mode number.. 74 3.6 Convergence of resonant slot length as a function of the number of modes. 75 4.1 A shielded stripline, uniform in the y direction with multi-layered substrate and superstrate................................ 79 4.2 Effective relative dielectric constant for shielded microstrip on alumina (a = b = 250 mils, W = 25 mils, c, = 9.7 ) compared to measurements [19] and Touchstone [20]............................ 87 X111iii

4.3 C: aracteristic impedance for shielded microstrip on alumina (a = b = 250 nills, V = 25 mils, Gr = 9.7 ) compared to Totuchstone.......... 4.4 MNulti-mode propagation constants for the even modes in shielded microstrip (a = b = 12.7 mm, h = 1.27 mm, w =.635 mm (, = 8.875)... 89 4.5 Stripline field distribution for the dominant propagating mode........ 90 4.6 Nhicrostrip field distribution for the dominant propagating mode..... 91 4.7 Suspended microstrip field distribution for the dominant propagating mode. 92 4.8 Fitlline field distribution for the dominant mode............... 97 4.9 Suspended fiilline field distribution for the dominant mode...... 98 4.10 A grid of vertical wires in rectangular waveguide............. 100 4.11 Equivalent circuit for single post in rectangular waveguide.......... 103 4.12 Mlagnitude and phase of reflection coefficient for a single conducting post in rectangular waveguide. Point match Method of Moments (MoM) results compared to the data of Marcuvitz [48] (A/a = 1.4)............. 104 4.13 Behavior of element values for the equivalent circuit given by Marcuvitz (48] (A/a = 1.4)............................ 105 4.14 Input impedance as a function of post diameter with a matched load port. Post diameter increases in the counter-clockwise direction from d/a =.005 to d/a =.25 in.005 steps, demonstrating the transition of the input impedance from inductive to capacitive.................... 106 4.15 Shunt susceptance behavior for up to seven wires placed evenly spaced across waveguide cross section........................ 107 4.16 Reflection coefficient measurements for a single post at various offsets from the centerline compared to predictions from the simplified model. Wire diameters (d =.025,.033,.039 in.) increase in the counter-clockwise direction....................................... 109 4.17 Reflection coefficient measurements for pairs of wires at various offsets from the centerline compared to predicted values. Wire diameters (d =.025,.033,.039 in.) increase in the counter-clockwise direction....... 110 4.18 Measured reflection coefficients for centered wire grids of equal spacing and increasing number. Wire diameters (d =.02.5,.033,.039 in.) increase in the counter-clock;wise direction....................... I 1 1 xiv

4.19 M.leasured and predicted reflection coeffcients for wire grids witli a siingle pin missing in the sequence. Wire diameters (d =.025,.033..0339 in increase in the counter-clockwise direction..................... 5.1 Geometry of basic coupler.............. I 5 5.2 Measurement of location of current peak relative to the slot for staindi ng wave calculation........ i 16 5.3 Photograph of coupler fixture assembly................. l l 5.4 Effect of the line stub length (1) on S21 and S11 magnitudes (s = 0, L, = 0.25 inches and f = 12.0 GHz)...................... 119 5.5 Equivalent circuit for a 2-port coupler.....................'20 5.6 Effect of the slot length on S21 and Sl1 magnitudes (s = 0, l = 0.049 inclies and f = 18.0 GHz)........................... 21 5.7 Comparison of theory and experiment for S21 magnitudes with s = 0, I = 0. 115 in., L, = 0.250 inches. The error bar indicates the influence on the high frequency corner of a ~.005 inch change in stub lengtl........ 1''2 5.8 Comparison of experiment and theory for S21 magnitudes with s = 0, l = 0.080 in., L, = 0.250 inches, illustrating the control of the high frequency corner with the line stub length by comparison to Figure 5.7........... 12.1 5.9 Experiment and theory for S21 magnitudes with s = 0,1 = 0.110 in., L, = 0.153 inches, showing the effect of shortening the slot length..... 125 5.10 Comparison showing the effect of line separation with I = 0.115 in., L, = 0.250 in. and s = 0.125 inches........................... 125 5.11 The effect of line separation with shorter line stub length; I = 0.080 in., L, = 0.250 in. and s = 0.125 inches...................... 126 6.1 Strip-fed cavity-backed slots with coupler-fed and series-fed corporate feed arrangements.................................. 129 6.2 Two cases for application of the Reciprocity Theorem............ 130 6.3 Normalized resistance as a function of slot offset compared to Shavit's measurements [67................................. 133 6.4 Resonant length as a function of slot offset compared to measurements [67]. 134 xv

6.5 Phliotograph of slot assembly pieces in various stages of fabrication.... 136 6.6 Uncompensated peak normalized resistance as a function of slot offset compared to measurements for uncovered slots.................. 137 6.7 Predicted resonant length as a function of slot offset for uncovered slots. Crosses indicate actual test piece dimensions to be compared with the frequency curve corresponding to the measured resonant frequencies listed in the table................................... 139 6.8 Sensitivity of resonant length to cavity length dimension'a'........ 140 6.9 Compensated peak normalized resistance as a function of slot offset compared to measurements for uncovered slots. 142 6.10 Compensated peak normalized resistance as a function of slot offset compared to measqurements for covered slots................. 143 7.1 Striplille excitation of waveguide-coupled slots.............. 148 7.2 Typical variation of impedance as a function of slot length for a conventional'thin' slot with slot offset as a parameter............... 149 7.3 Variation of impedance as a function of slot lengtli witli the difference in top and bottom slot lengths as a parameter................. 149 7.4 Slot coupling (resistance) and resonant length dependence on offset and waveguide length (Lw)............................. 150 7.5 Apparent bandwidth comparison for identical slots versus slots with different lengths based on an interpretation of the relationship of slot length to wavelength......................................151 7.6 Apparent bandwidth comparison for identical slots versus slots with different lengths based on calculations at selected frequencies......... 152 8.1 T-Bar fed slot geometry.......................... 161 8.2 Two cases for application of the Reciprocity Theorem for the T-bar fed case. 162 B.1 The derivative of an arbitrary discontinuous function............. 177 xvi

LIST OF TABLES Table 6.1 Measured resonant frequencies (GHz) and predicted resonant slot lengths for uncovered slots with compensated cavity length. (Deviation from actual slot length listed in parenthesis in percent.)............... 142 6.2 NMeasured resonant frequencies (GHz) and predicted resonant slot lengths for covered slots with compensated cavity length. (Deviation from the actual slot length listed in parenthesis in percent.).............. 144 xvii

LIST OF APPENDICES Appendix A. Vector Wave Function Expansions and Relations................ 165 B. On the Use of Vector Potential Functions for the Derivation of Green's Functions...................................... 171 B.1 Potential Theory Solutions for Electric Currents........... 171 B.2 Field Behavior- Electric Current................... 179 B.3 Potential Theory Solutions for Magnetic Currents.......... 181 B.4 Field Behavior - Magnetic Current.................. 183 B.5 Partitioning............................... 184 C. On the Use of Vector Potentials with tile Method of Scattering Superposition 186 xviii

CHAPTER I INTRODUCTION 1.1 Motivation and Background Thle usc of slots in niicrowave circuits can be traced back at least as far as the 1940's to the research efforts associated with World War 11. This is particularly true in antenna designs wlhere slots lhavc been used extensively as the primary radiating elements, but also in feed networks as couplers. With the advent of monolithic microwave circuit technology and also for more traditional construction methods, transitions from microstrip to slotline are also becoming increasingly important in the design of microwave and millimeter-wave circuit elements. Two such transitions can be combined to form interconnects between lines and by using lines on opposite sides of the slot plane, a vertical transition can be made without the use of via-holes. The basic structure, in a variety of forms, has a wide range of applications to both broadband and narrowband connections and can be used as a building block for interconnecrs [35), phase shifters and inverters 130], directional couplers [80], filters [503, and many other microwave components 12]. Whether used as a coupling element between guided wave structures or to free space, both applications require accurate analysis and design tools in order to minimize costly, time-consuming empirical techniques and rework. Many papers have been published on the design of slotted waveguide antennas over the past forty-five years including "classics" by Stevenson (71] and Oliner (57], among

2 otliers [51, 4, 5]. Elliot and lhis associates have contributed many important works ol the analysis and design of slotted rectangular waveguides and arrays (22 23. 5S. 70. 40i. Miore recently, slots wliich are fed by microstrip or stripline have received attention (59. 62. 6G, due to advantages in cost, size, weighit and conformability, among others. The development of variations and new analytical techniques is ongoing [16, 8, 17, 87]. Numerous investigators have also presented approximate analytical techniques to chiaracterize these types of structures with applications to circuit elements [12, 49, 50, 41. 21. For example, a quasi-static analysis has been provided in [89], however, this may not be sufficient, particularly for higher frequencies where end effects and hlighier order mode coupling become more significant. A more recent paper [66] presents a transmission line analysis with excellent results, however, similar shortcomings would be expected. IHybrid methods which combine two-dimensional full-wave analysis with transmission line tlieorv as in [56], should certainly extend the validity of such models, but still may not account for all discontinuity effects. Exact methods for microstrip-to-slotline transitions hliave onvly recently begun to appear, such as the case reported in (91] which has applications to open structures. For couplers, practical considerations suggest that a shielding structure will almost always be present. In fact, in many cases, a shielding structure must be introduced in order to reduce crosstalk and to control undesirable coupling to othler structures in the package (for example, DC control lines in phase shifters have been known to unexpectedly become part of the microwave circuit). For antennas, cavity-backed slots have been used for similar reasons; for example, to reduce internal mutual coupling on the feed network side of an array. This often both simplifies the design process and improves the achievable performance. For instance, it has been shown that the internal isolation of the slots in phase steerable antennas can significantly improve scanning performance [47].

3 For thtese reasons, the slots studied lhere will he enclosed by a cavity at least on one side. The dimensions of the cavity can lave a strong itifluence onl thte electrical behavior of the slot. in some cases, tlhese dimensions are used to conitrol thie slot clharacteristics and in others they are used to suppress undesirable effects. In practice, slot antennas are often covered by a protective dielectric sheet. The capability to handle tlhese cases will also be included in the analysis discussed here by employing the exact Green's function for an infinite covered ground plane. Tlis functiont will be evaluated by a combinationl of nlumerical and analytical techniques as described in [36], however, the details will not be reiterated here. Also included is the capabilit to miodel multi-layered substrates and superstrates which is becoming increasingly important for monolitlic circuits and for systems which may combine many circuit functioiis through three dimensional initegration over multiple layers. An example of a hypothetical strticture whiich employs numerous transitions of tile type to be analyzed is illustrated in Figure 1.1. Shlown is a slot antenna array loosely based on a conventional waveguide fed slot array. Conventional arrays typically use waveguideto-waveguide slot coupling to feed slotted waveguide branch lines. IHere we assume the individual slots are fed by microstrip-to-microstrip slot couplers as illustrated by the lowest three layers of Figure 1.1. Each radiating slot is internally isolated by a cavity which may contain active devices for power generation for example. The feed network may contain additional active sources coupled by combinations of microstrip and slotlines and mav be built on multilayered substrates for integration of additional antenna functions, such as phase control, frequency conversion, detection, etc. This example illustrates just one application with features drawn from topics being considered in the current literature which also verifies the need for more accurate and advanced analysis and design tools for the types of structures to be considered.

g - ~- - / ~R adOrne EL~6~ttng S\OtS i Cavy t tso.aoUO CGenerat _Aodu\es sCng ots coupvint ircults Bias Co Cks f tures* ~~~~~~~~fXt4ue Tuts

1.2 General Description of Analysis Approach For coupling through slots, tlhe distribution (shape, amplitude aniid pliasc) of tlhe Field in the slot is vital to the determination of the slot coupling behavior. For radiating slots. this distribution has been investigated by Elliott [23] and others [59. 67. 701 by treatiing tlle case of an isolated slot in an infinite ground plane. With the Green's function for a half space and the fields in rectangular waveguide, they were able to write an integral eqiiatioln. The equation was solved by the Method of Moments to determine the field distribution. Subsequently, the scattering parameters of the slot were found by an applicationi of the Reciprocity Theorem. The same approach will be used here for radiating slots, referred to lhere as the lReaction Theorem Method' or'Reaction Method'; although the presence of the strip alnd. especially for microstrip, the inhomogeneous filling of the rectangular cross section precludes tile use of simple rectangular waveguide field solutions. In previous works (67], tile presence of the strip has been neglected by assuming a similarity between the strip modes alld empty waveguide modes. This not only places certain restrictions on the position of the strip between the ground planes but also does not take into account the influence of tle strip on the fields near the slot. Here, the formulation will be generalized to include not only the case of stripline where the cross section is homogeneously filled, but also the possibility of multiple layer substrates and superstrates, allowing microstrip or more complex structures to be analyzed. This will require the solution to the two-dimensional'waveguide' problem in addition to the three-dimensional'cavity' solution which is the primary focus of this work. The relative position of the strip within the shield will also be unrestricted whlich has not been the case in some of the previous works. For couplers, the same shielded strip substrate/superstrate capability will be included. However, in this case, an alternative method will be used to extract the coupling parame

ters. This will be referred to as the'Standing W'ave Method' and is based on a niore detailed modelling of the strip current. The scattering parameters are extracted from the positions of peaks in thie strip current standing wave pattern, evaluated for even and odd excitations of the structure. Although the Reciprocity Method is more general, the Standing Wave Method is somewhat simpler and therefore may be more convenient. In many cases the tliickness of the ground plane is very small and the slot can be considered infinitesimally thin. However, for both types elements it may be necessary to make a'thick' slot for structural purposes or heat dissipation, for example. In fact, we have found that this feature may be used to another advantage which is to broaden the operating bandwidth of thle element. These cases can be treated by the same techniques and will be itncluded in the formulation. Highly accurate models are needed to reliably design these types of elements. Therefore, a full-wave space-domain integral equation approach solved by the Method of Moments will be used as opposed to quasi-static, modified quasi-static dispersion analysis, or other similarly limited techniques. This approach is preferred since we can find exact Green's functions for the structures described which account for all possible electromagnetic interactions. It also allows application of these techniques to problems without restriction as to the size of dimensions relative to wavelength. 1.3 Integral Equations and Notation In this section, appropriate integral equations will be derived. One of the primary purposes of this section is to introduce the notation and conventions used throughout the text and some of the fundamental equations and relationships which will be needed.

1.3.1 Dyadic Green's Functions for Physical Quantities It is well known that time harmonic electromagnetic fields must satisfy Nlaaxwell s Equations ( eJw' time convention assumed and suppressed throughout): V x E=-jpH (1.!) Vx H = j+wE (1.2 where the constitutive relations D = (E and B = pH have been assumed. Also. J is defined in terms of the movement of electric charge with time by thle Continuity relation: J = -jwp (1.-3) Understanding these to be the governing independent equations, by taking the divergence of Equations (1.1) and (1.2), it is found that the fields also obey p V. E=- (1.4) V -H= (1.5) That the fields must satisfy the Helmholtz wave equation is readily derived by taking the curl of Equation (1.1) and substituting (1.2) into the result, yielding V x V x E- k2 E = -jWJ (1.6) where k2 = w2pc. Similarly, it can be shown that H must satisfy VxVx H-k2H=VxJ (1.7) Dyadic Green's functions can be introduced to represent the solutions to these equations for infinitesimal current sources. For instance, eJ will be used to represent the solution to Equation (1.6) in the form -eJ = Ez). + I) + (

Here,'(z) is tile field resulting from an: directed current source jr) = -(R-R') 19 and so forth for the y and z components so that v x v x C,eJ - k2Ge, = I~ (R - R') (1.10) where l6(R - R') = -jwp[Ji) + JY) + Ji)] (1.11) Likewise, for the magnetic fields, the dyadic Green's function must satisfy V X V X,mJ - k:2mJ = V x [(6(R- JI')] (1.12) where GmJ represents C. ni = -jwPs[ ()i + HlY + T(Z)i]Z (1.13) From Equation (1.10) and (1.12) it is seen that GeJ is related to GmJ by V X GeJ = Gmi (1.14) V x G m = s1(R - R') + k2C ej (1.15) The terminology and notation which we use to refer to these functions is that GeJ is the dyadic Green's function of the electric field type (subscript'e') for an electric current (subscript'J'). Similarly, GmJ represents the dyadic Green's function of the magnetic field type (subscript'm') for an electric current. 1.3.2 Dyadic Green's Functions for "Dual" Quantities The Duality Principle entails the proposal of a system of equations where V x E =-j.T H- K (1.16) V x H = jwE (1.17)

; representing a fictitious magnetic current. This current in introduced as a matter of convenience by which we can represent tangential electric fields in terms of equivalent magnetic currents according to the Equivalence Principle. We then can take advantage of the duality between these and the "physical" quantity equations to generate solutions to the equations [32, pp.98-116]. The Helmholtz wave equations for this system are V xVx H - kH = -jw (1.1) V x V x E - k2E = -V2 x Ki (1.19) Using the representation GmK = [Hi+ ZHMf + TY ) ] (1.20) Gnil< satisfies v x V x GmK- _k2C mK = i1s(R - R') (1.21) where the inhomogeneous term represents fb(R - R') = -jwE[R iz+ Ky + (')i (1.22) with -') (1.23) and so forth for the y and i components. Similarly, GeK satisfies V x V x V eK - k2eK = V x I6(R - R') (1.24) where GeK represents eK = jw4(f + Y + E (1.25) Here it is evident that V X mK = CeK (1.26) V xGeK = I6(R- R') + k2 GmK (1.27)

The notation and terminology are the same as before withl the magnetic currelnt represenited by the subscript'K'. 1.3.3 Integral Solutions to Helmholtz Equations using Dyadic Green's Functions Once Gej, GmJ, GeK and GmK are known for a particular geometry and set of bounrdary conditions, it is possible to find the corresponding field quantity for any distribution of electric or magnetic current. The relationship to be used can be derived from tile vectordyadic Green's second identity: JJJP[*.vx v x -(V x V x P).Q]dV — Jj{fi xVxP.Q=+(ix P) -xQ}dS (1.28) To find the integral solution to V x V x E- k2 E = -jw (1.29) we use the electric dyadic Green's function with an electric current source, GeJ. By letting P = E and Q = e.l in Equation (1.28), we find that E( R) = -j J(i'),( Ri', R)dV' - |s(t[p x V x E( R )] * eJ (, R.) +[x.) x P(:)) -' x e,,j( R', R) } dS' (1.30) Note that in the process of deriving this result, the notation of R and Rt' has been interchanged in keeping with our conventional usage of R' as the source position vector and R as the observation position vector. (Primes throughout will be used to indicate source coordinates.) Using the vector-dyadic identity of triple products: a.(b x c) = -. (a x c) = (( x b) - (1.31)

Equation (1.30) can be re-writteii as E( R) -,= j J(R,) Ge( R, R)d +jL{[V' E(R')[fi x Gej(R' R) -[nx E(R')] [V' x Ge( R', R)] dS' (1.32) In our application, GeJ is used in a cavity with impedance walls representing the source layer. It is assumed that thle region outside this layer is bounded by the impedance walls and a surface which requlires either the radiation condition or the Dirichlet condition for the E and Ge functions so that tile [nx ] terms in Equation (1.32) evaluate to zero on the boundary. We also assume that this region contains no sources. If such is the case, the surface integrals of Equation (1.32) disappear and we are left with E( R) = w // Ge,(. R) R J( (R')dV' (1.33) Here, the symmetry property of the dyadic Green's function: GeJ( R, R') = [GeJ( R', R). (1.34) has been applied ('T' indicating'the transpose'), which can be shown using dyadic-dyadic Green's second identity as outlined in [76] under the assumptions stated above. We can find the integral solution to V xVx -k2 = xJ (1.35) by returning to Equation (1.28) which with P = H and Q = GmJ, reduces to H(t) = JJf,mJ(R, R) J( R')dV (1.36) using assumptions similar to those for the E fields with the appropriate radiation or Neumann boundary conditions enclosing the impedance walls. Alternatively, Equation (1.3G)

can be obtained directly from Equation (1.33) by using HR R)= — v x E( R) 1.37 and V x Ge = GmJ (1.38) The procedure for the magnetic currents is exactly the same resulting in H( R) = -jwJG mK( R, RI) KR( R') dV' (1.39) and E( R) - JCeK( RX PR) v K( Re)dV' (1L40) 1.4 Formulation by Application of Boundary Conditions We now derive a set of integral equations by enforcing the boundary conditions for tle problem. To begin, we replace the slot openings in each region by a tangential, conductorbacked equivalent magnetic current in accordance with the Equivalence Principle as illustrated in Figure 1.2 (all conducting walls will be assumed to have perfect conductivity). The problem is thereby separated into independent regions, coupled together by the magnetic currents as shown in the figure which also illustrates the treatment of finite slot thickness. Since K = -fi x E, using the same current on either side of the-slot openiiigs enforces the continuity of the electric field in the slot. In addition to the boundary conditions at the cavity walls, slot walls, ground plane surface and dielectric interfaces which will be satisfied by finding the appropriate Green's functions, we must also satisfy the following boundary conditions: XrF'" = t (141)

13 Cover Slot z c z =d' me' a % -K Z=C E KL cav Cavity z=d Strip Impldancc Boundary Figure 1.2: Cavity layout and equivalent magnetic currents for a radiatinig slot. on the microstrip; fX nt _ ix Rst0L = (1.42) at the slot opening to the cavity; and n x 1Lotu _ i x Ti = X (1.43) at the slot opening in the dielectric covered plane. The E function is a vector describing the source and is normally set to zero except when a gap generator is used to excite the line1. The X1 function is used in Equation (1.42) when the slot is excited internally2, or in Equation (1.43) when the source is external3. When E is set identically to zero, Equation (1.41) enforces the boundary condition that the tangential electric field is to be zero on the strip. Equations (1.42) and (1.43)enforce continuity of the tangential magnetic fields over the slot openings when'i is set to zero. The'u' and'L' subscripts will be ulsed'The gap generator is used only for the coupler problem as a mechanism for even and odd excitation of the lines. 2The right hand side will be set to the incident h field of the dominant strip mode for the Reaction Method formulation. 3The right hand side can be set to an incident plane wave field for the analysis of the slot as a scatter-r or receiving antenna, however, this problem will not be discussed further in this work.

throughout to indicate tile upper and lower slot openings for the general case of a'tliick' slot. The electric field, E", in the interior of the cavity can be written as an integral in terms of the electric and magnetic currents in the cavity using the relationships outlined in Section 1.3.3: n= GeJ:,nt, Js dS' -' GeK,int - KL dS' (1 144 ) strip lot L Similarly, H anywhere in the cavity is given by nt G mJ,int - Js dS' -jwEc I n G mK.int L dS' (1.45) strip J lotJ L For a thick slot as shown, the H field in the slot can be written as H.o, =jWEJ I0 mKlot. KL dS'-WjEIf CmK.slo'r Ku dS' (1.46) _lot L lotu The external H field is given by - = jwEd L mKext Ku dS' ( 1.47) slot u Using these expressions in the boundary conditions given by Equations (1.41-1.43), we arrive at the integral equations for the problem: -)Wi c iJjn X eJtnt~ JJ dS' hn X f K,int KL dS' = ~ (1.48) X mJit J dS' - - j~ fi x ml.in' KL dS' trip mJ9nlotlldS JWdS -"j.W.Jj1 fix C'mK.Aot KL dS' + jw.J 1o fi X mKlotL Ku dS'= 7 (1.49) loft fL oto 09'- j ILL n x.mKet' gu dS' = 7 (1.50)

where the superscripts on the dyads imply evaluation of the dyad at thle indicated lont ot Ttiroughout this work, Equation (1.48) and similar types will be referred to as an electric field integral equation (EFIE) since it is derived from a boundary conditioii onl thie electric field. Similarly, both Equations (1.49) and (1.50) are magnetic field integral equations (MFIEs). Since each of Equations (1.48-1.50) involve the [fix ] term, from here on it will ~be dropped witli the understanding that only the tangential components of tlhe various dvads are used. Equations (1.48-1.50) can then be written: j(i.LC G eI.int' J dS' - eK.int KL dS' = ~ (1.51) strip slot L f f IL -[ [1L =slot 1 G mJ.int Js dS' - mKIint + fsGc ml<.slot K L ds + jw, CmK.slot f 2 Ku dS = ~ (1. 2) ) 1W ] /rn m K.slot ['U + dm i =' (1.53) where all the terms involving i are excluded4. The unknown currents J,, KL and Kft can have only tangential components on each of their respective structures, we thlerefore have six scalar unknowns. Since only the tangential components of j are used, Equations ( 1.511.53) represent six scalar integral equations which are sufficient to solve for the unknown currents. For convenience of notation, we redefine Equations (1.51-1.53) as: fJ )J,.[dS'+ KL dS'+ ) L5 ( 4) 1t30 ) d35+| L u Kju dS'= X= (1.56)'Throughout most of this work we will consistently use i = i where n, will be orthogonal to the currents involved. Therefore, no integral equation or Green's function evaluation will involve the i components which will subsequently be ignored.

16 where =(3I) -— =t rip G ()= e C1J.int! =-( Geje.gl ( 1.38) =(31) = (1.9) G G-eK.in,t c(31) = slot L( G=(35) -jS KlotL G(= JWSG mK.sloG (1.61) -(53) = jWT,lotu = jWS nmKslot (1.62).tlotu =,tot ] ( j) j mK.slot + (dGm K.extj (1.63) For the radiating slot problem, it remains to find the various dyadic Green's function components, solve the equations for the unk;nown currents, and interpret the results to obtain parameters which characterize the slot's electrical behavior and properties. The coupler problem differs only in the final equation. Equation (1.56) which involves the Green's function for the half-space is replaced by an EFIE as folows. The field on the upper slot is now contained by another cavity which may also have a conducting strip as illustrated in Figure 1.3. Under these circumstances, we then have the following integral equations: j. JL dS' K KL dS' = (1.64) i/Jd i/ J [,+.o ] K lopt *lot ff J ~.JudS' + j,~,|| fUmK + slotmK' K dS' dl,p m',L. dS'- aslo G6 +-./,,JU dS' + f| KP - vU dS' = ~ (1.65) -, o. if., [,Lo,,oot The loumJ'sridS' + and K h ida wloer Ku dS' riPtr ~lott T s c atsu hrip trtpu soU u) ff ff'*T, t+ipy oly~~~~~~~~~~~~~~1.T The subscripts v and r, here~~ Jnict slothe Uh orei soitdwtrte\pe

17 Impedance Boundary Strip z = d' Upper Cavity z=C EC v,L L ~Ca~~V,L Lower Cavity z=d Strip Impedance Boundary Figure 1.3: Electric and equivalent magnetic currents for microstrip-to-microstrip coupler with a'thick' slot. lower cavity strip or slot opening.

CHAPTER II DYADIC GREEN'S FUNCTIONS In this chapter, tile dyadic Green's functions needed to evaluate the integral equations are derived. Tile methlod used in all cases is a'field expansion method' using Vector Wave Functions (VW'Fs) as opposed to the perhaps more conventional,'vector potential method'. Some of the Green's functions derived will not be needed in the later analysis but are included in this chapter for completeness. The use of vector wave functions with dyadic analysis is presented as an alternative. It has the advantage of producing the complete dyadic Green's function in one solution. The process also may involve a reduced number of simultaneous algebraic equations which must be solved for unknown coefficients as compared to the vector potential method. Tile disadvantage of the approach is that it requires dyadic analysis which may be unfamiliar, although it is quite straightforward. The field expansion approach using VFWs has been extensively detailed by Tai over an extended period of time 172]-[79]. Nevertheless, the method is not widely employed which may be due to several factors including: 1. Early development of the technique involved the use of the vector wave functions designated L, M and N. As will be shown, the M and N functions have clear, physical interpretations, however, the interpretation of the L functions is somewhat obscure

and its handling, at times, somewhat difficult. Consequently, the approach may have been avoided. 2. An oversight in [72] but corrected in [77], stirred up some controversyl wlich may have caused some to avoid the method (see [11) also for details and a list of references). The current method is more mature, having evolved to a stage where the previous difficulties have been eliminated. Problems can now be solved in a methodical and straightforward manner with no difficulties in physical interpretation (the need for the L functions can be avoided). 2.1 Impedance Boundary Conditions for Layered Structures It can be shown that a plane wave in a homogeneous region exhlibits a constant wave impedance defined by the ratio of a component of the electric field to an orthogonal component of the magnetic field, both transverse to a given direction 16, p.142]. A'plane wave expansion' of the field is convenient in many problems due to this property. For many canonical structures, the expansion itself is unnecessary since wave impedances can be derived directly. For instance, modal wave impedances for homogeneously filled rectangular waveguides are well known. In somewhat more complex structures such as those treated here, the boundaries are still always planar owing to the rectangular geometry. As a result, wave impedance surfaces can be chosen to conform to the boundaries and tlle wave impedance concept becomes a vehicle through which the boundary conditions can be applied in a simple way. The dyadic Green's functions for all of the structures treated here will be derived using this approach. As the Equivalence Principle states, the fields in a given layer depend only on the fields at the boundaries and internal sources. We therefore can derive the Green's function for the source layer alone with the other layers represented by impedance

boundary conditions applied at the layer interfaces. In this way the Greents functioii catll be generalized to represent any iiumber of layers above or below tile source layer. Tile approach greatly simplifies the analysis of the total structure by allowing the fields to be found first for the source layer alone. Once they are found, expressions for the fields in the remaining layers can be immediately written in terms of the homogeneous solutions by matching tangential components on the boundaries. As in [13], this process is greatly facilitated by expanding the solutions in terms of'Longitudinal-Section Electric' (LSE) and'Longitudinal-Section Magnetic' (LSM) fields because the field matching procedure can then be done on a one-to-one basis. (An individual mode on one side of an interface matches an identical mode on the other side exactly, with an appropriate coefficient.) The impedance boundary conditions, as used here, are not to be confused with the approximate impedance boundary conditions discussed in [63. 64]. Both usages may be exact under certain circumstances. The present usage is in the context of the modal impedances of various structures and is exact under the assumption of perfectly conducting walls where applicable. For example, the approach is exact for a closed, perfectly-conducting rectangular cavity with uniform side walls. A counter-example is a cavity with perfectly conducting side walls, but which is open on one end: terminating the open end with the impedance of free-space as proposed by some, is not exact since this condition is not exact. For all structures studied in this work, the geometries are such that the representation is exact to the extent that perfectly conducting walls can be assumed. In our structures, the impedance boundary conditions require the fields to satisfy j-E Fo Heti cret.hibc(2.1) For electric currents, this becomes m - -- or _ = jWolr (2.2) O * mJ O, mJ

21 Similarly for the equivalent magnetic current Green's functions, the impedance boundar" conditions are; i. GeK zi GeK JT7 -Jwmr77 or V e (2.3i Y*CmK X CeK 8 For simpler notation, rTi will be the wave impedance associated with the LSE modes and Tim wiUl be used for the LSM modes. Furthermore, ir and 91m will denote normalizatioln to the intrinsic modal impedance in the layer and are defined as ki rlei k,i = i (2.4 imi = (2.5) kzs where i is an index associated with the ith layer. We can then evaluate the impedance conditions using transmission line analysis alnd wave impedances for the various layers as illustrated in Figure 2.1. The impedances on a........................... -nu Figure 2.1: Impedance boundary condition representation of a multi-layered structure. given layer's upper boundary are found by the transmission line equation k(..i = +,e(j- +() )tan k,(i-1)l(i-)] (2.6) Ymi -- ~ik..i-). [m..i_l)+ jtaltk..i_.,'(i_..]'

where the index i is iterated from the top layer through successive lower layers to the layer of interest with 1, the thickness of thle i'h layer. (We have assumed that, = 1 throughout this work.) Similarly, for the lower layers, since the wave impedance is negative. 1e, -- kz' [1e('+"-jtankz"+')l('+"] (2.8) k,+i) e+ - J(i+l) tan kz(,+l)l(,+l igket| - ____ [n(a+Il) - j tan kz(i+1)1(1+l)] E,+lk() j Tn(,+l) -tankZ(+,) (+,1) (2.9) - (,+i;k..i [ - jOm(,+l)tan k I+ where here the iteration proceeds from the lowest layer upwards. 2.2 Dyadic Green's Functions for an Infinite Covered Half Space We begin with the derivation of the Green's function for an infinite covered half-space. The method of solution closely parallels that of the otlier cases, except for the boundary conditions, so that by coverinig tlhis case in greater detail some of the steps for the later cases may be omitted. For completeness, we also include the solutions for electric currents in this section, although this function is not needed for the characterization of the slot. It is, however, widely used in the analysis of open microstrip and microstrip patch antennas. 2.2.1 Magnetic Current The dyadic Green's functions for the slot problems use a magnetic current ( K) as the source. They are the solutions to the dyadic Helmholtz equations: V X V X GmK-k2PCmK = I6(RR- R') (2.10) V x V x GeK - k2,eK = V X 6(R - R') (2.11) The key to eliminating the need for the L vector wave functions, and thus simplifying the analysis, is in the choice of which of Equations (2.10) or (2.11) to solve first. The functions L, M and N form a complete set of solutions to the homogeneous equation

23 V x V x F - 02 F = O. They are defined in the general forms L = v 1 (2.12) M = V x:i, =-V x [N (2.13) 1 1 = -V x V x =- x M (2.14) K; K where' is a scalar function solution to the equation V2' + r2,I = 0 chosen to satisfy the boundary conditions of the problem;:i is a unit vector called the'piloting vector'; and t is the separation constant c2 = k] + k2 + k2. Following the Ohm-Rayleigh method as described by Tai [721, we can find the particular solutions by expanding the right hand sides of (2.10-2.11) in terms of the eigenfunctions L, M and N with unknown vector coefficients; deriving the values of the coefficients using the orthogonality properties of the VWFs; expanding the dyadic in terms of the same functions with scalar coefficients; and enforcing the equations by performing the derivative operations. From Equations (2.12-2.14) it can be seen that only L can have a non-zero divergence and since the right hand side of (2.11) has no divergence, the L function is not needed in its solution. We therefore find the solution of (2.11) first. It can also be shown that GmK and GeK are related by V XmK = eK (2.15) V x CeK = I6(R-') + k2C mK (2.16) so that CmK can be found from (2.16) once,eK is known. This is the essence of the method described in [77]. As with the vector potential method (see Appendix C), we begin as if the space were infinite and homogeneous. We therefore expand the field in terms of VWF's for free space defined by = e-J(k'+Lky+") (2.17)

24 To simplify later application of the boundary conditions, we choose the piloliiig vector to be Z. The orthogonality properties of these functions are then jI IM(k,ky, kz) -N(-k'-kt -k )k d' = dO ((218) f/f M(k, ky, kz) k)M( -k', -k,I -k' ) dV = IL N(kx,ky, k,) N(-kI,-kI,-k')dV = (2 )3(k2 + k2 )6(kr - kr ).(k I-k' )6(k,z - k) (2.19) where the volume of integration corresponds to thle entire space. To find eK we first let V x [l6(R - d)] jJdJ dkdkdz [Kl(k t(kzk,) )A+ N(kr,ky., k:)1A(220) By taking the anterior scalar product of Equation (2.20) with M(-k. -k,l. -k'.) and N(-k/, -k, -k') respectively, and integrating throughout V, we can determinine the unknown vector coefficients A and B through the orthogonality properties. The results are A L(7k, -k,-kz),(2.21) A (27r)3(k2 + k)(2.21) 2M (-k., -kif? -k..) (2.22) (2r)3(k) + k2) in which the primed functions are defined with respect to (z', y', z'), the site of the source at R = R'. Thus, V bI6aRR"'1 = 1 00 00 D00 dk,,dkdk. *[M(ks ksokz )N(-kss-kyl-ks)+ N(k,ky,k..)Ml(-k.k-k,-kz)1 (2.23) Now we let eK = (2. ) 3 JJJ 1 dkdd *[a1P(k, kJ,:)N(k -M, -k, -k,) + bN(k=,k, k,)M'(-k, -k, -z)I (2.24)

25 withl a and b unknown scalar coefficients, and substitute into Equation (2.11) yielding 1 1 K2 k2 [k2 - (k2 kk2 - ) The integration with respect to k, in Equation (2.24) can now be carried out in closed form by applying Cauchy's theorem: GeK -82 I k(2 + 2) M(k)N (Tk ) + N(k )(tkz) > z' (2.26) where k = k2 - k- k2 (note that n becomes k). As indicated, the top sign applies whien z > z' and the bottom sign when z < z'. This condition is a result of requiring the solution to satify the radiation condition at infinity which determines wlhether the contour of integration is closed in the upp-: or lower half-plane. Also, from here on it will be understood that the primed functions, M' and N', have -k=, -ky or -k, arguments, unless indicated otherwise. We now can write GmK by performing the operations indicated by Equation (2.1G). This can be done almost by inspection using the relations between M and N from Equations (2.13) and (2.14) except for the discontinuity which occurs at z = z'. As in [TT1, it can be shown that these relations apply but an additional term is needed to account for the discontinuity at the source; specifically, VX C;eK jA~ Jo'p dkTdk [ k(k + dk M) (~k,)M (Tk.) + N(~kz)-N ( Fk V X eK =-8s2, k,(k2+k, ) +I,,_ R-,') z<> z' (2.27) where the transverse idemfactor, It = i: + yy, appears in this case as a result of thle combination of the choice of the piloting vector z and the partitioning of the z dependence. Notice that the singular terms come from the second derivative of f(z) =:'L('< > ( 228) ~fk, lz-z) Z 2_

or the first derivative of -ekk,(z-) z g(z) = f(): (2.29 e.k(z-z') < z' which is e-ka(z-z') z > z Og( z ) _ a f ( Z ) = 2 ___z - =Z2 / k- 2jk,6(z - z) (2.30) eJk,(z-z') z < This term has sometimes been overlooked in the potential function method as well, as discussed in Appendix B. Thus, Equation (2.16) leads to GmK = 1~~6'~ ~ f~fO 00 dk,dk1, GmK j (. ) zi2J R -y(k2+.2 8,r2 -oor kz(kA+kl) [M(ikz)M'(:Ft k) + N(+k,)N'(Tk,) )] Nz' (2.31) We now divide the infinite space into layers surrounding the source point above and below. The layered structure can be represented by impedance boundary conditions for the source layer as previously described and as illustrated in Figure 2.2. We then apply.,...-............. 2. z=c IrlujFigure 2.2: Layered infinite space as represented by impedance boundary conditions. the method of scattering superposition to this case by letting - e p) =-(S) GeK = ~eK + GeK (2 12

PeK is the particular solution found above for the infinite space which may be referred to as the primary term, and Gel( - tile secondary or scattered fields - is made up of solutions to the homogeneous equation as follows: S) k / f'00 dkzdkv Gel = 8ir2 II oo kz(k+2 + k2) M [(k,) A+ + M(-k,) A + N(k,) B + N(-k,) B-j (2.33) where A+, AB+ and B are unknown vector coefficients to be determined. The physical interpretation of this procedure is that the ~ coefficients represent the waves traveling in the fi directions as a result of reflections, i.e., scattering, from the interfaces. In evaluating the boundary conditions, it is also useful to find Gml= through V x Gel< = k2GmK which is the source-free version of Equation (2.16). Hence, =(S) j f t dkzdky GmK = 8,2 J k.(k2_ + k2) *N(kz)A,+ N(-k1)e A-M Bk+ + Nu + M B + M(-kz) - (2.34) Note that additional boundary conditions need only be imposed at the newly introduced interfaces and not at the source, since the primary fields satisfy all boundary conditions at the source and the secondary terms are continuous there. Applying the boundary conditions at the top and bottom of the source layer, we derive the following set of algebraic equations for the unknown coefficients: (ieu 1)C-jte A+ - (i?eu + l)ecJ' CA = -(ieu - l)e j kc' (-kz) (2.35) (cL- l) d A+ -(NcL + 1)eCJkd = (ieL + l)eCkjdN(k ) (2.36) and (tirn - l)ekwc jB+ + (i7u + l)ciekc B' = -(rm - l)eijvclM(- kz) (2.37) (mtmL- l)e)ksPd B+ + (fmL + l)e3kld B- = -(fmL + l)efrldM'(kd J- (2.38)

Here the'u' and'L' notation indicates the upper and lower interface conditions respectuvei. These four equations decouple into two pairs due to a judicious clhoice of tle piloting vector in the VWFs. Because the piloting vector was chosen as the unit vector normal to the layer interfaces, the M and N VWFs appearing in GmK correspond to the LS, and LSE modes of the structure respectively. (In contrast, it should be noted that M- and N correspond to the LSE and LSM modes, respectively, when they appear in Gel<.) As is well known, the LSE and LSM modes are decoupled on the interfaces, that is, the tangential components of an individual mode on the interface can be matched by an identical mode in the adjacent layer, therefore the coefficients are decoupled. That there are only four equations is a result of the fact that the field in a given layer depends only on the field in the adjacent layers. The solutions are easily found to be _ + eekvd [IV eU - )('eL + 1)e-Jkc' (-k.k) + (f/eu + )(ONeL + l)eJk' CN(kz)] (ke;U + l)(fieL - 1)e_k,(c-d) - (f/eU - l)(fL + l)e-j*,(c-d) (2.39) cjkevc {[(feU - l)(7eL- 1)e-jk"dN'( —kz) + (eUv- 1)(feL + 1)e'7dNR'(kz)] (McU + 1)(LeL l)e)k,(c'd) - (f/eU- l)(_cL + l)e)jk&(c-d) (2.40) _ + jkd [(fimU - l)(0nmL + 1)ewcM'(- k2) - (f/mu + L)(00mL+ I)e "M'Mc(kz)] =(rnU + I)(f/mL - 1)ejkv(cd) - (fmu - )(00mL + 1)eiJk1,(cd) (2.41) B= e ek*c [(1mru - )(U - nL l-)ej;kdM'(-k) - (7mu - l)(mL + l)e2k'dM'(k)I (fmU + l)(mL - l)ejkv(c-d) _ (fmU - 1)(00mL + 1)ejk(c-d) (2.42) With some algebraic manipulation and use of the relations found in Appendix A, we now can write GeK = k, |(k3 + k2)

29 |( [ieuKei[kz(z - c) - j Ol[kt ( - c)I] [7ieL INkZ' - d)] - jSi-kz' - d)l (eU - eL) C kz (c - d) - ( L - 1)sin kz(c - d) [,m~uNo[kz(z-c)] + jNe[k,(z- c)]] [O,LM[k,(z' - d)] + j[- (z' ] [7,LNo[k(:z - d)) +j jNe[kz(z - d)]] [iulMo[k:(Z' - c)] + jKlt[-k(z' - )] (imu - rmnL)COskz(c- d) - j(tmUmL - 1) sin k,(c- d) for:' (2.43) where the Mo and Ne functions are defined by cos(kz )e-J)(~k+kVV)'I:(k,) = (2.44) sin(k,z )e-J(kz+ kI) This expression contains all components of the dyadic Green's functioii selparated into LSE and LSM modes. Note that our convention will be to denote even and odd trigonometric dependence by the substripts'e' and'o'. In order to avoid any ambiguity when these subscripts are used, the corresponding k., k, or kz arguments will be shown explicitly in the same order, sometimes followed by other arguments as appropriate (see Appendix A). The magnetic field type is given by CmK zz6(R o dkrdk, 7- d) k -,oMo[k ( - ) + k ) + j4k(z c) (i,rmU - hmI) cos k,(c - d) - j((i,mU ImL. - I)sin k,(c- d)

30, [t,e.,e'[kz( - - c)] - JNo(k,(z - )] [lK[k(' d) - t- -( - d)1 ] T[7eLNe[kz(z - d)] - jNo[kz(z - d)]] [ieu [k,(z - c)] - iNo-k2(z - c)j] (,U - eL, ) cos k,L(c - d) - j(kUf L, - 1) sin k,(c - d) for z' (2.45) When applied to the case of a slot on an infinite, perfectly conducting ground plane, t,L is set to zero. For a sinigle dielectric cover layer, on the upper interface we set k,eu = k (2.46) ko T)mU = Er (2.47) corresponding to the normalized impedance boundary conditions for free-space above the slab. For a slot in the ground plane with its axis along 2, we take the Z > z' terms wlich results ill 1 1 f dk.,dk,,_jllr5(t-r')+)~(yY)I,m,- ='4~r2 J oo k,z(k2 + k2) [,k,sin k(z - d)+jkcos k,( - d) k2k2 k, sin k,(z - d) + jk, cos kk,(z - d)+ \ rk,, cos k,( d)+jk sin k,( - d)J k kcos k,(c - d) + jk, sin k,(c - d) (2.48) By transforming the spectral integrals to a cylindrical coordinate system the double ilitegrations can be replaced by a single radial integral on recognition of the integral representations of Bessel functions in the angular variable. Through some very tedious algebraic manipulations, the result can be transformed from the present form, which separates the LSE and LSM modes, to a hybrid form which can be compared to the result in Appendix C or with previously published forms 138, 36]; although not necessary for numerical evaluation.

2.2.2 Electric Current For the case of the electric current source, the steps are exactly the samne as in tlho preceding section, except here the Helmholtz equations take the forms: v x v x eJ e = (R - =') (2.49 v x v x GmJ - k2GmJ = V x l6(R - R-') (2. 0) where now we solve for CmJ first. Due to the similarity to the previous set of Helmhioltz equations and because of the way the VWFs have been defined - particularly the symmetry introduced in the curl relationships - the solution proceeds with identical equations but with the following notational replacements: M' - N (2.51) GeK o CmJ (2.52) CmK e cJ (2.53) The process is similar to replacements made under the guidance of the Duality Principle. however, it is important to note the difference. The replacements dictated by the Duality Principle alone would result in functions which satify'dual' boundary conditions, i.e., the electric field dyadics would satisfy the Neumann rather than the Dirichlet conditions on the conducting boundaries [13, pp. 29-39]. By replacing the M and N functions with each other, the true boundary conditions remain valid since these functions are complementary with respect to these boundary conditions. Although it would be interesting to more generally state and define this process under a heading such as say, the'Similarity Principle' the development and proof is beyond the intended scope of this work, however, we will us it repeatedly.

32 The final result then is jk tx /' dkrdk, 2mJ = J0 o kz(k2 + k2) [mU Mio[;kz(z - c)] + jM,i[k(z - c)]] [L [kz(z' - d) + d)I] 1mLMo[kz..( - d)] + jMe[k(z - d)]J [.mUo[kz(Z' - c)] + jN[-k)]] ) (imU -,mL) COS kz(c - d) - i(f mUmrnL 1) sin k,(c - d) feu&Ne[kz,(- -)] - jNo[k,y( - c)l] [keLM'[kt(' - d) - jMo[-k,(z' - d)] [77LN[kz(z- - d), - jNo[kv( - d)],[euMM[k(z' - C)]- iM'o-k,(z' - c)] (UeU -,c ) cos kt(C - d) - j(iieUiL - 1) sin k,(c - d) for z z' (2.54) Gej can be found by substituting the coefficients into equations similar to Equations (2.322.34), but also can be derived from GmJ directly, applying Equation (1.15) with special care in performing the derivatives at the source discontinuity. The final expression is G 1 W R R"a( - -/ 00 dk.dk,.=k2ZZi R-R)+ 402J oo J0,ks(k2 + k2) [TUMeN[kz(z - c)] - Mo[k[k(z d)] - jM[-k(z' - d)I] |t[ flo[kz(z - d)] - jMo[k,(z - d)]] [7M Ue[kz(Z' - c)] + jMN[-k(z' - c)1] ) (iJmU -'mL) cos k.(e - d) - j(femUfL - 1) sin k,(c - d) for z>z' (2'7n

2.3 Dyadic Green's Functions for Layer Filled Rectangular Waveguides The electric and magnetic dyadic Green's functions for an electric currelit source in rectangular waveguide are needed for the radiating slot problem. As will be shown later. the electric type is used in an integral equation approach to solve for the propagation constants of the structure. Once a propagation constant has been determined for a selected mode, both the electric and magnetic fields on the entire cross-section are required to apply the Reaction Method to the three-dimensional cavity problem. The magnetic current case is not needed in the main body of this work but will be discussed briefly in Chapter IV. As in [79], the solution for the multi-layered waveguide problem is built upon tihe solution for the parallel plate problem. The parallel plate waveguide solution has already been obtained in the previous section if we set iiu = iL = O on the source layer boundaries. However, although the parallel plate solution we need is based on VWVFs defined with respect to a piloting vector i as was used above, the desired planes for the parallel plates are defined by z = 0 and z = a which do not correspond to the impedance boundaries used previously. Therefore, the parallel plate solution will first be derived based on the appropriate conducting planes followed by the layered rectangular waveguide solution. 2.3.1 The Parallel Plate Green's Function A parallel plate waveguide shown in Figure 2.3 is formed by bounding walls at z = O and: = a filled with a uniform dielectric material represented by the wavenumber k,, where i will ultimately represent the ih layer of a waveguide containing a source. We now define the functions Mo(k,) = V x'o(k,)i = V x [sin(m,z/a)ej(-,+ka)]iI (2.56) Ne,(k:) =- -vx x [=cos(m,ra/,)ej,-t."+e't ~ (2.57) K K

34 x x-a ki k (x'.z') x -=O z Figure 2.3: Parallel plate waveguide coordinate system. which satisfy V x V x F- I2 = O0 where K = /k2 + k2+ k? and the boundary condition i x F=0 (i.e., i x E=0) (2.58) at z = 0 and z = a. The wavenumbers kv and k, are two coltitiuously distributed eigenvalues and ks = mxr/a where m is an integer including nz = 0 for Ne. The orthogonality properties of these functions are JJM(m,k, kz). No(m',k, k' )dl' = 0 (2.59) for any combination of even and odd functions and for any two sets of eigenvalues (m, k,,, k) and (m', kl,k'). The volume of integration corresponds to the entire space inside the parallel plate waveguide. The normalization constants of these functions are stated by the following relations: ///| Me(m k,, k, ) -,(m',, -k' dV - JJf N,(m,kz,,k). Ne(m',-k;, -k')dV 0 m# nm' {=1+6m)22r2a(k~+k~)6(ky....k~,)6(k <(2.60) (1 + 6m)2r2a(k~2 + k2)1(1,, - k)r(k, - k') m = m' = 0, 1,2,...

35 IlL Mo(mrky, k:) t0(m', -k, -k dV = Jj N'o(m, k, kz) N'o(m', -k', -k' ) dV 0 m m "i' =t~~~~~~~~~~~~ ~~~~(2.61) 2r2a(k2 + k2)6(kv - k )6(k - k' ) m$ n'= 1, 2,.. where 6, is the Kronecker delta function defined by 6m = m= 0O m$0 To find GCn.j we first let V x [16( R - R') = / / dk,dk, [0o(m, ky, k) Ao + Ne(m, ky, kz) Be] (2.62) 0o oo m=O By taking the anterior scalar product of Equation (2.62) with Mo(m', -kg, -k') and Ne(m', -k,, -k' ) respectively, and integrating throughout V, we can determine the vector coefficients A0 and Be through the orthogonaLity properties. They are,-o (2 - 6, )KNo(m,_-k,, -k2) o = 4r2a(k+k) (2.63) 4r2a(k2 + k2) Be (2mKM(,y,z (2.64) In Equations (2.63) and (2.64) the primed functions are defined witl respect to (x', y', z'), the location of the source. Although for m = 0 the function N' vanishes, m = 0 is included, as implied by the factor (2 - 6,) in Equation (2.63), to put it in a form similar to Equation (2.64). Substituting Equations (2.63) and (2.64) into (2.62) we obtain V x [l6(R- R')] = J,ddk 4r2a(k2 ) [Mo(m, ky, k,)No(m, -k, -k,) + Ne(m, ky, k,)M'(m, -ky, -k)] (2.65) Now we let 0mJ = | | " (26m)K J:L:dk mdki,o 4w.2a(k) + k k) [aI.o(m, k~,,kb)Nb(m, -k,, -ks)+ bN,(m, rk,,k)l7',,(m, -k,,k:.)j (~266)

36 Substituting Equations (2.65) and (2.66) into (1.12), and making use thle rclatiolis gi\vn by Equations (2.13) and (2.14), one finds a = b = b = (2.67) as before. The integration with respect to k, can be carried out in closed form bV applyillg the residue theorem together with the radiation condition yielding oo j(2 - 6,)ki GmJ d= 00 m=O y [Mo(m, ky,~kz)No(m,-kT,: kz) + N,(m, k,, ~kz)M(m, -, kT)] z z' (2.,S) where kz = x/k?- k2 k_. Again, the top sign applies to z > z' and the bottom sign to z < z'. Now applying Equation (1.15), again through use of the relationships given by Equations (2.13) and (2.14), and talking into account the discontinuity at the source, we can write e R - R, dk j(2- 6, GeJ = -2 i6( R )- J - dk z 4ak(k2+ k2) co m ---—' M[l(m, ky,,~kl)M(m, -k,,:k,) + NO(m, k,, ~k,)N'(m, -k,, Tk,)] z Z' (2.69) The solutions for the magnetic current follow the same procedure yielding GmK = - i6(R- R ) Jdk A2 - 26) ~ [Mo(m,k, ~k,) a(m.) + N(m, k,, ik,)N',(m, -k,,:k,)] Zc Z' (2.70) eK do, kE 4wrak,(k2 + k2) [iM(m, k., ~k~)N`,(m, -kr,,k.) + No(m, ky, ik)MO'(m, -k,,k,)] z z' (2.71) Note again that this can also be obtained from the previous case by simply replacing CeJ -: GmK, CmJ:- CeK and M. N.

37 2.3.2 Rectangular Waveguide with Electric Currents We now construct the dyadic Green's function for the source laver of a multi-lavered rectangular waveguide (see Figure 2.1) by using the parallel plate Green's function solutions above and the method of scattering superposition. Let: -(P) (S) Gmj = mj + G mJ (2. 2) =(P) =(S) where G mj is the parallel plate solution and G m is defined as 00dk 4+rak(k2 +k) [N.(m k,, kz)A + + e(m,,-kz) - + M (m, k, k.) B+ + Mo(m, k,-k) B-] (2. 3) representing fields which are scattered from the dielectric layer interfaces located at z = c and z = d. The unknown vector coefficients, +, A- B+ and B-, call be found by applying the upper (i7') and lower (77L) impedance boundary conditions for the layer. This produces two pairs of equations for the unknown coefficients: (rieu - 1)e-jk'.+ - (iOu + 1)ceJi = - 1 )e- jk'M'(m, -k,,k) (2.74) (?eL - l)e-jksd + (ieL + 1)e"kjd A- = (ieL + 1)eCkdMe(m, -ky,-kz) (2.75) and (4,~m - 1)e-jkc~ B+ + (imn + l)ejkC B- = -(,mu, - 1)e-jk'cN'(m, -ky, k=) (2.76) (imL - l)e-ikad B+ + (m, L + l)ejk' aB = -(mL + l)eik'dNo(m, -k, -kz) (2.77) Notice that these equations are identical to Equations (2.33-2.35) with the replacements: M'(~kI ) = M',(m, -k,, ~k) (2.78) N'(~k.) fNo(m,-k,,::k,) (2.79) This feature is characteristic of all of the solutions we will be dealing with, and should be expected since the functions differ only in whether the functional dependence is exponentia

or trigonometric. As a result, the solutions are identical to Equations (2.39-2.42) with thie corresponding notational substitutions. Algebraic manipulation of Equations (2.68,2.72) and (2.73) anid use of tihe relations given in Appendix A then leads to _ fdk j(2 - 6m)k; GmJ dkm O 2rak(kk + k2) M( o[mu; m, k(z - c)J o[,mL; m, k(z' - d)]. Mo[.mL; m, k.(z - d)]J j[imu; m, k,(z' - c)] (,mU -,mL) cos k,(c- d) - j(tmUtmL -'1)sin k,(c- d) X elrue[; m, k.(z - c)] M,'[T),L; m, k,(z' - d)] e (e[ieL; m, k.(z - d)J Alf[ieu; m, k.(z' - c)] (Nev - eL)COs k,(c - d) - j(icUieL - 1)sin k,(c - d) for:>:' (2.80) where we have now defined the new operator functions, Me and Ne, for the sake of compact notation: M0o(7;;a] = 77Mo[o] + jMoe4[ ] (2.81) fM,.[;ca] = 7Mee[l - jM,[a] (2.82) Ao[;ct;a] = /No[4aJ + jN,[c{] (2.83) Mdv7i; al = 1t1,,o[a - ji,[o] (2.84) Since a here is [m, k1(z - c)] and m is associated with k, we recognize that the'e' and'o' subscripts in this case imply trigonometric functions of z and z. Also recall that the primed functions use -k,. We can find GeJ as we did in the parallel plate case, by performing the derivatives

39 indicated by Equation (1.15): = 1 rc~~o j(26- m,) G, = - ( R - R)+ |_d;; C 2rak;z(O;2- + k) Me[jjeu; m, k,(z - c)] M)[74CL; m, (' - d) MelIcL; m, kz(z - d)] M[f,eu; m, kz(Z' - C)] (e - cL)cCOSk,(C - b) - j(ilOeUr.L - 1) sin k,(c- b) /o[lmnu; rn, k,(z - c)1 A'[,)mL; m, kZ(z' - d)] ~Ao{wmL; m, k,(z - d)J Ao[i,nmu; m, k,(z' - c)] (,mU - Om~L) COS kz(c- b) j(-imUimL - 1) sin k(c - b) for z> z' (2.85) Both GeJ and CG mJ involve a spectral integral which can be reduced by Cauchy's Theorem, once the impedance functions are specified. 2.3.3 Rectangular Waveguide with Magnetic Currents The magnetic current cases can be found through the same method or'Similarity Principle' substittutions (section 2.2.2) to obtain I.' / oo j(2 - b, ) G mK =-7.iz( R-R ) + dc, 2xrak,(k2 + k2) M4o[0m,u; m, k,(z- c)l MX,[i,t,; m, k,(z' - d)l MO[,imL; mr, k(z - d)J [iimu; m, kz(z' - c)l O(mU - mL) cos kz(c - b) - j(i,miU4mL - 1) sin k,(c - b) ( e[ m kz; m, k,(z' - d)] A'e[TCL; m, k,(t - d)J] [ri,u; m,(z' - c)] ('JeU - 2eL)cos kz(c - b) - j(ieUfe.L - l)sin k,(c - b) J for z z' (2.86)

40 and 00 f j(2 - 6,m)kGeIK = 1J00 dk, ZE 2lrak,(k2 + k2) te(e[cu; m, kr(z -c)] Al[cL; m, k2(z' - d)] Me[l,eL; m, k,(z - d)] A',[(u; m, k,(z' - c)] (CTeU - ~eL) cos kz(c -b)) - j(eUcL 1 ) sin k,(c- b) |Roiimu; m, k(z - c)] Mlo)mL; m, kz(z -d )] m (zo[mL. d),] AA,nu; m, k(z(z' - c) (,mU -,mL) COS k.(c - b) - j(0mCJ,mL - 1) sin k (c - b) for z z' (2.87) 2.4 Dyadic Green's Functions for Layer Filled Rectangular Cavities The scattering superposition approach can now be applied to the result of the previous sectiont directly to obtain tihe dyadic Greeni's functions for the cavity problem by initroducing conducting walls at y = O and y = b and applying scattering superposition to the ~y directed waves. As an alternative, one can take a somewhat simpler approach by first deriving the dyadic Green's function for a waveguide with its axis along the i direction (k, becomes nw/b). Then scattering superposition is applied along the i direction with impedance boundary conditions to obtain the result for the cavity (see also (72, 75, 79]). This will be the approach demonstrated here since the intermediate Green's functions will also be needed in Chapter IV. 2.4.1 Homogeneously Filled Rectangular Waveguide: TE and TM Modes As with the use of the half-space solution as a building-block for the previous solutions, the preceding modal representations of layered rectangular waveguides are not in a convenient form for the formation of the cavity solution. Again, the VWFFs there are defined

41 with the normal to the layer interfaces, resulting in the LSE and LSNI mode repreenta tion. W'hat is more convenient here is tlhe solution for a homogeneously fillcd rectanguiar waveguide e>xpressed in terms of VWYFs defined to represent the modes Transverse-Electric = (W) (TE) and Transverse-Magnetic (TM) to the waveguide axis. We again will first find Gmj =(W) from which Gej will follow. = (W) Gmj must satisfy the wave equation: V X V x GmJ kaGm = Vx (-R )I (2.88) To construct the solution we will need the vector wave functions satisfying tile Neumann boundary conditions which are: Moo(k.,ky) = V x'Io(kt,ky)i (2.89) N,(kx,k,) = X V X,,(k.k,k,)i (2.90) where cos k:z cos ky'k e (k,s k~) = e1" ek (2.91 ) sin k,z sin k,y with ks = km = mr/a and k, =k = = n/b. Following the Ohm-Rayleigh method as before, we expand the source term as v x I[6(R - R)J =j |dk, F [oksZy) A + ~ (k, k| )B(2.92) The coefficients A and B are found from the properties of the vector wave functions to be A- (2 ) - 6n)x (k, k),-k,)) (2.93) wab(k2 + kj)N( k -= (2-..,.,)_ B = ab(k+) k2) M(k1. k,V*,-kg) (2.94) where the Kronecker delta function 6, is equal to 1 for m = O or n = O and 0 otherwise (the case where both m = O and n = O is the trivial, zero field solution for this-wase) Thus

42 Equation (2.92) can be written as =00 00 (2 - mn) v x [ lb(,R- R')1 = k. E,-b( dk,,, + J~ O rab(k + k) m=O n=O *[Moo(k, ky, I)0oo(k, ky,, -k,) + NS,(k,, k,)l.' (kk, -et) (2.95! To findn,J we let Gmi J = 0dk Z,:rab(k2 +k2) m=O n=O m n a[alMoo(r ky kz,, )NOo(k_,, k,,- k,) + bNe(k,, k,k, k)M.(k, k,,-k,z)] (2.96) Substituting into Equationl (2.88) we find as usual a = b = (2.97).2 _ kC so that = (W) = 00'0 (2 - bmn) _ G mi'-', dk o,o ~ rab(k2 + k2) jK2 - [.Mooi(k, ky, k:)Noo(kz, ky,-k:) + Nee(k,, ky, k,)M'(k,, k -kz; )j (2.98) The Fourier integral can be evalutated in closed form by means of contoutr integration and the radiation condition leading to =(W) jkj(2 - mn) Gmj n==o abk,( k2 + k2 ) [Moo(kr,. ks,:~kz)Noo(k,l k,l:kz) + Nee(kr, ky, ~kz)M'(k., k, F:kz ) for z,z' (2.99) where k2 = k2 (k2 + k). Using Equation (1.15): (W) I - ~w~ co ( _2 -mn. ) 3eJ = Ok2"' ( ),n=O,=o abk,(k + k2) M [o.(k, k,l ~kz)N'(k., k,k, Fk,) + ee(k,k, ~k, )M.(k,, ky,:Fk,) for z z'. (2 1()()

43 2.4.2 Cavity with Opposing Impedance Walls: Electric Current WVe now introduce impedance walls at z = c and z = d where tile added bound-rconditions given by Equation (2.1) are to be enforced. Using the method of scattering superposition we let = (W) =(S) Gmj Gmj GmJ (2.101) where = (S) jki(2- 6mn) Gmj ZZ ~=o abkz(k2 + k2) [Moo(k, ky, kz) A+ + Moo(k,k,-k) A- + Ne(k, k,,, ) B + N+((k., k,) Bk] (2. 102) Evaluating the boundary conditions given by Equations (2.1) we get the same system of equations as (2.33-2.3.5) except this time with M'( ~k ) * Me( (k,. k,,:kz) (2 03 N'(~k,) _ Noo(kz,k, ~k, ) (2.104) Again we already have the solutions for this set by changing the notation of the VW\Fs. With some algebraic manipulation and use of the relations in Appendix A the results are q i0 2jk-(2 - 6bm) mJ m=o n=o bkZ(k2 + k2) ((-c) A[,mL;k, k k,(z' - d)] Moo[lmu; k, ky,,(2-C)l -'1 V I' |Moo[lmL; kr, kY, k,(z - d)] R lWmu; k5, ky, k(z' - c)1 (,mu -,,mL ) Cos k(c - d) - j(fi,imL - 1) sin k,(c - d) (fee[ieu; kxzk,, k,(z - c)] M[e k,; ky k,kz(z - d)] Cf.M..[;;k,. 11k, t(z d)] M-'.[i2.u;k.,k.,k*(z' - c)) r (ieu l lLr)CLcoskz(c - d) - j(i?eufieL - 1)sink(c - d) for iz'iz' (2.10'

and Gej- + E bk(k +k2) m=O n=O a — (Mee[i7eu; kri kyz,ik,(z - c)] Me[eu; kr, ky, kz(z' - d)] Iee[(eL; k:, k,, k,( ] - d)] -4[)eu;k,, ky,k,(z' - c)) (cU eL) COS k.(c - d) j(t eU eL - 1) sin k,(c - d) RJoo[mu; kr, ky, kz(z - c)] )0[r,lmL; kz, ky, kz(z' - d)l.,oo[mL; k., k,,, k,(z - d)] C-o[Imu; k:,,k, k,(z' - c)] (imu - Qm,L) cos k.(C - d) - ( j(mmU1mL - 1)sin k,(c - d) for z> z' (2.106) The operator functions are defined by- the relations M~o[77;T = 11oooA [o] +jMo r (2.107) Mee[l/; a = ilMeee[al - jMeeo.[ (2.108):00[ool7;c] = t)ooot[a] + j0ooe.[Q] (2.109) Jl;{]; a{ = Neee.,[o} - jNeeokl (2.110) 2.4.3 TE and TM Modes in Homogeneously Filled Rectangular Waveguide and Cavities with Opposing Impedance Walls: Magnetic Currents To model the slots, we also need the cavity dyadic Green's functions of both types for magnetic currents. The derivation could follow the previous case explicitly, however, because of the way the functions have been defined, we can take advantage of the symmetry of the equations and write the solution by making simple notational replacements. (The only exception is the treatment of the (2 - 6,,) term which here is expanded as (2 - 6m)(2 - 6,) since the m = n = O case may produce non-zero field components.) Using this approach we can write the TE and TM solutions for magnetic currents in rectangular

45 waveguide as =(W) oo o jk:(2 - b,)(2 - 6n) GeK On=o abk,(k2 + k2): m=O n=O f* [Mee(k~k, ~y k,)N,(k,k, kFk) + 4oo(k, ky,~kz)Noo(kZ, ky, Fk) < (2 111) M* oo(k, ky:!~k )Moo(k k, k,) + <(k,r k,,:k k, ),: The cavity solutions are Cx aE, s0, 2jki(2 - bm)(2 - bn) GeK = _m0, akbk(2 + k2) (.I,[f:u; k= ky, k,(z - c)] kee[?,L; k,,l kz(Z' - d)]' Mee[.c; k, kt, k,(z - d)] ),A[,lu; k,. ky, k,(z' - c)] (cu - j cL) Cs k,('( -d)- j(ie,u,icL - l)sin k,(c- d), oof[imu; k., k,. k,(z - c)] 1M' [m L; k5, k,, kz(z' - d)] Ajoo([mnL; k:, k,, k,(z - d)] M1 [i7mu; k,, k, k(z'- c) (rmu -,mL) cos k,(c - d) - j(rmUorml - 1) sin kz(c - d) for z<z' (2.113) and CmK = 2ii(R - R))+ 2(2-)(2 - ).loo[Oimu; kx, ky, k,(z - c)] M) [o,L;k, kt, kg, k( z' - d)] jt Aoo([fmL; k5, k, k,(z - d)l AMoo{[mu; kw, k,, k.(z' - c)] (vmu - imLr) cos k(C - d) - j(),ifmU,L - 1) sin kd(c - d) for zkz (2- -, d)]

46 WVe now have derived and specified thle dyadic Green's functions for all types of sitr-c tures to be treated in tlis work. The integral equations are thierefore fully definied anld tIe task remaining is to solve for the unknown currents in each case and interpret tlhe results.

CHAPTER III METHOD OF MOMENTS FORMULATION A general methlodology for application to the various problems treated will be presented in this chapter for the case where two components of current will be allowed on bothl the strips and slots. Later, we will restrict our attention to strips and slots which are narrow so that only the lotngitudinal component of current need be considered. This assumption is sufficient to yield accurate results for the experimental cases to be used for verification. and thus simplifies the numerical implementation without loss of generality. The solutions to thle presented integral equations can be found by choosing basis functions to approximate the various currents. The error in the approximation is minimized by applying the well known Method of Moments, resulting in highly accurate representations of the currents from which the electrical belhavior of the structures can be deduced. Furthermore, the method of moments formulation will be discussed in the context of thle radiating slot problem only, since this problem contains all the essential elements of the coupler problem as well. 3.1 Definition of Coordinate Systems and Basis Functions Let us expand the current on the strip in the following manner. We first define a strip-fixed coordinate system as illustrated in Figure 3.1. The currents on the strip in this

48 ya xo x Figure 3.1: Cavity (z, y) and Strip-fixed (v,v) Coordinate System. coordinate system can now be expanded as J. = J + +J'i (3.1) Jv = (,,(v) (v, )IuV, (3.2) J. = (m) l (3.3) where 4 are piecewise sinusoidal basis functions defined by 1 sin kb(a- aq- I) for -lo < a - ca q < O0 ~(a) = sinki. (3.4) Sl sin kb(aq+I - a) for O < a - < c 1o and 1, is half the subsection length defined by Il = [aq+ -- Xq-l]/2 (3.5) The subscript q is an index identifying basis functions at various points along the strips and slots. Actually, kb will always be chosen so that kblo < r/2, making the function basically a triangular pulse. This way, because the basis functions overlap, the current will essentially be approximated by piecewise'linear' segments between sample points (see t321 for an introduction to the method of moments and basis functions). The sampling rate is determined by field phenomena, phase resolution requirements or numerical limitations. as

49 will be sliowii later. Typicalvly, tile samprling rate will be at least 20 samples per wavelength. often much higher, so we generally set kb k, which is more than sufficient to'linearize the basis functions. The motivation for the sinusoidal dependence is to simplifv later integrations and evaluation of the resulting functions. The'I function will be either a'Maxwellian' distribution or a'pulse' basis function defined as'IMfaxweIlhan(13) = it(=) = 1o'-(~/' )~ I - Bql < 1a (3.6) The Maxwellian function is often used since it closely approximates the true solution for narrow strips or slots (.52, 88]. TLhese expansions are furthler illustrated by Figure 3.2 where the sinusoidal functions are exaggerated for clarity. Longitudinal -UZ piecewise sinusoidal L/2 -W/2 W/2 Maxwellian Transvese -L/2 pulse - recuangular L12 -W/2 wn/2 Figure 3.2: Current Expansion Functions. In this work, we will deal exclusively with strips and slots which are narrow with respect to wavelength so that only one basis function will be used to represent the narrow dimension. For wider structures, rooftop functions are commonly used, involving similar overlapping basis functions in the direction of each component of current, but using the

50 pulse basis function for the transverse dependence. Because our strips and slots are n;arrow. we will typically only model the longitudinal component of current, however. botli components are discussed here for generality and to form a basis for future efforts. The primary motivation for the use of piecewise functions is that thev are very efficient in terms of changing strip or slot lengths as opposed to entire domain basis functions. The strip-fixed system is related to the cavity coordinate system by v = (z - o,)cosO + (y - y,)sinO (3.7) l = -(z - z0) sin X + (y - yo) cos4 (3.8) We also define a slot-fixed coordinate system as illustrated in Figure 3.3 where y 0 sO x Figure 3.3: Slot-fixed Coordinate System ((,(). ( = (z - so) cose + (y - to) sin E (3.9) = -(t - so) sin + (y - to)cos6 (3.10) The slot currents are now written as Ku = Kc(C + (1- i l 1I,

where the'L' subscript represents the slot opening to the cavity and thle'u' subscript represents the slot interfacing with the dielectric cover. Assuming similar expansions for these currents, KL( = ( (( ()I;L(j (3.13) KU( = P(s)Zt((.j)Vu(j (3.14) KL~ = V,(() u VL~ (3.15).7 h'u( = 4 Z([) VU(' (3.16) Equations (1.54-1.56) can be written in matrix form Zil Z12 Y13 Y14 0 0 I, 0 z21 Z22 Y23 Y24 0 0 I, c Z31 Z32 Y33 Y34 Y35 Y36 VtLC (3.17) Z41 Z42 Y43 Y44 Y45 Y46 V L(J 7 o 0 Y53 Ys4 YSS Y56 VU( O 0 Y63 Y64 Ys Y s VUvt X: where each term in bold face is a submatrix described by integrals such as zi= 1' Z Z2 = J Gl")(z, y, z', y'),P(v1)((v.)dSj Z2 =1 aJ G(,)(: y, z', y')y(uj )), (vt)d S Y/, = Gl 1c3 (, Y,', Y,)'(G)0((j)dS,

52 or in a more compact form as (I ) G(12) C(13) G(14) GV GUA/ ) G( 0 0, I'(v), )fl, C(21) C(22) C(23) C(24) 0 v0(t)I, c C(31) (32) G(33) G(34) G(35) C(36) ((j)VL( i dZ G(41) G2)( G(4) G(r5) C(G6) l((2 )4(0)VL4, dy (u -V 4( t( "( "'(( o 0 C(53) G(54) G(55) G(56) |I ((A)VU( | o0 0 (. G(3)5) G(G) G((')('.VU) 4)Vu, (3.18) The functions on the right hand sides of Equation (3.17) and (3.18) are discussed in Chapter I and are further defined in section 3.2. Note that S, is the jth'source' segment of the corresponding strip or slot. We also have a triply mixed coordinate system which must be accommodated using the transformations given by Equations (3.7-3.10). For example, computation of C( 13) involves both G(13) and C(:3). In addition, the derived Green's functions are in the cavity-fixed coordinate system. Consequently, all terms must be transformed to a common coordinate system before the integrations can be performed. Treatment of these integrations and manipulation of the various G terms will be discussed in a later section. Tile evaluation of the elements of the matrix at the positions where tlle boundary conditions are being imposed has not yet been discussed. Following the conventional method of moments formulation, we introduce a weighting function and impose an inner product to be evaluated at each subsection on the strips or slots. The inner product is defined as (as) =f|Ja*b dS (3.19) where a = i, the weighting function, and b will be the vectors represented by Equations (1.54-1.56). Note that the elements of Equation (1.54) are to be evaluated on the strip as indicated by (1.57) and (1.58). Similarly, Equation (1.SS) is evaluated on the lower

53 slot as indicated by Equations (1.59-1.61)and Equation (l.56)on the upper slot as impilied by (1.62) and (1.63). Following Galerkin's method we can choose iL to have the same forni as the basis functions used in the expansions of the currents. Thus, for Equation (1.54) tb = T(v),(v)i + %r(v),(v)t i3.20) Similarly, for Equations (1.55) and (1.56) t, = D (c)$(()~ + P((),(c) (3.21) The elements of the matrix in Equation (3.17) are then clhanged to Z,= If /S q(v),,(v)Z,,dS, Z'2'= fsj (V)4(v)Z'2dS, -y3v = Jf'(4)4(()'/ajdS, where the i index represents the'field' point integration locations. For certain cases we will use point matching on the slots driven by considerations in the evaluation of the Sommerfeld integrals of the half-space Green's function. In tlhis case the weight function for the transverse dependence of the longitudinal component of magnetic current on the slot becomes a delta function resulting in the evaluation of the field at a point at the center of the slot. 3.2 Excitation Models The final element to be discussed is the excitation vector represented by the right hand side of Equation (3.17). As mentioned previously (Section 1.3), thle' terms represent

non-zero values in the excitation vector corresponding to gap generator locations on!he strips. Likewise, the Eti terms vill be non-zero for the incident H field excitation used witih tihe Reaction Method. In this case, the incident field must be weighlted the same way as the left hand side so that the right hand side terms become 7Hg =|fJ?i2D.nc dSi (3.22) For the gap generators, the corresponding field for the gap subsection can be designated E9. Application of Galerkin's method then, results in the integral JJ. = || Eg dS, (3.23) on the right hand side of Equation (3.17). E, is an unknown caused by a source at that location on the microstrip. If we assume'E = E,,i,, then we can set - O in the second row of Equation (1..54). In most cases, we can arbitrarily set the integral of (3.23) so that 4 of the first row of (3. 17) becomes a zero column vector except for one element corresponding to the position of the gap generator of the form E=[ 0 10... 0 O T (3.24) where T denotes the transposition operator. Setting the magnitude of E, is arbitrary because the Standing Wave Method used w: h the gap generators uses relative interpretations of the resulting current, not absolute quantities. For certain problems, however, we need the use of the gap generator model to determine absolute values of current on the lines. If we set the field in the gap such that the voltage over the gap given by Vo =- 9dl (3.2.5)

is equal to 1 Volt, thle input impedance at the feed point is given by Z,, = 1/19 to htr-: order. A similar model can be used for slots with a coaxial feed, as in [21, p.3GO for example, by replacing the coaxial feed with a current source on the slot. Before moving on, it is worthwhile to examine this process a little more closely in the context of some of the terminology and physical interpretations of gap generator models found in the literature. Restricting this discussion to one dimension, at the source our system of equations represents the enforcement of the boundary condition u wf(z) dz = K (3.26) ource where w is the weight function, f(z) is the field quantity and K; is the constant specified on the right hand side of the matrix equation row corresponding to the location of thle source. It is clear that the only non-zero contribution to the integral can occur on the domain of w for which wV is non-zero so that any physical interpretation is confined to that region. It should also be noted that there may be an infinite number of solutions f(z) satisfying this equation and that this equation does not force further constraints on what f(z) might be, i.e., the right hand side does not specify how f(x) behaves on a scale smaller than the domain over which w is non-zero. We then also have no basis for a physical interpretation which imagines the terminals of the source within this domain, but rather we should interpret the terminals to be at the domain's endpoints. Obviously, K will depend on the nature of w. The physical interpretation of the nature of the source then also depends on tw. For simple cases such as a pulse weight function, the delta-gap phlysical interpretation is appropriate since Equation (3.26) reduces to a form similar to Equation (3.25). The point match case, or delta-function source can be interpreted in the same manner by taking the limiting case of the pulse weight function, shrinking its width to an infinitesimal gap while keeping the area constant. For more complex weiglht functions, the physical interpretation is unclear except that we can consider the source to

56 be'distributed' over the domain of the weight function. The system of equations represented by Equation (3.17) is now fully specified. The unknown currents JS, EL and Ku can now be found by solving for the matrix elements and inverting the matrix. 3.3 Expansion of the Dyadic Green's Functions for the Cavity To evaluate the elements of the matrix, we need to expand the dyadic Green's functions into the components corresponding to the electric and magnetic currents which are in x-y planes. All of tile required termins canl be separated into trigoiiometric functions of I,z',y and y' multiplied by a complex coefficient which contains the z and z' dependence. The Green's functions can then be written in a condensed form as follows. Also, a constant complex coefficient C,, can be factored out which appears in all Green's functions for the cavity and is defined by 2j(2- 6,) Cmn =,,k + k (3.27) Cavity EFIE - Electric Currents J The electric field integral equation contribution of the electric currents involves the z-y components of GeJ which can be written in the form G eJxx = E Cm.n eJxx cos kmX sin k,ny cos kmx' sin kny' m=O n=O G eJyx = E Cmn G, eJyx sin km.z cos k,y cos k.z' sin kny' m=O n=O G exy = E E Cmn 5eJxy cos k,,Z sin kny sin Akx' cos kny' m=O n=O GeJyy = E E Cmn 5cJyy sin kmcosknysin km'coskny' (3.28) n=O n=O

57 where k, = k, = mxr/a, ky = k, = nr/b, and 9;eJxx = [kz2ee( (c ) + (k, ) c )Z (7 3,) 5eJyy = [k2 2ee(eIe) +( k( ) Zee (7m)] (3.30) 5'eJyx = 6eJxy = -kmkn [Zee(1le) - k)2,,,(iimM)] (.331) The z dependence of the Green's function is contained in the Z,, function defined as [(fU cos k,(z - c) - j sin k,(z - c)] [r'L cos k,(z' - d) - j sin k,(z' - d); Z(i) = [ cos kz(z - d) - j sin k(z - d)] [ cos k,(z' - c) - j sin k(z'- c)] (u - L) cos k-(c - d) - j(OUTL - 1) sin k,(c - d) for z>z' (3.32) Here again we use the'ee' subscript notation to imply the trigonometric dependence of tlhe function. Cavity EFIE - Magnetic Currents K The contribution of the magnetic currents involves the components of GeeK wliich can be written as GeKxx = ~ C,,n eKxx cOs k,,z sin k,y sin kmZ' cos kn,y' m —O n —O GeKyx = E E Cmn 5eKyx sin kmZ cos kny sin k,mz'cos k,y mO n-=O coo GeKxy = E ~ Cmn 5eKxy cos km sin k,y cos km' sin k,y' m=O n=O CeKyy = E ZCmn eKyy sinkmTC tkoskcos kmX sin kny (3.33) with jeKyx = -k, [knZeao(e) + k2 zo(i7m )] (3.34) QeKxy = ks [2kZ( + kZo(i) + k m)] (335) QeKxx = - 5eKyy = kmknkz [Zeo(i'e) + Zeo(m)]p (3;I,

58 The Z,, function is defined as [ u cos k,(z - c) - j sin k,(z - c)] ['iz sin k:(' - d) + j cos k,i -' - d)! tZo = [ilccos k(z - d) - jsin k(z-d)] (U sin k(z - c) +cosk:(: -c)]l (t7U - rL) cos k,(c - d) -.(k11U~ - )sill k(c - d) for c- -' (3.37) Cavity MFIE - Electric Currents J The magnetic field integral equation contribution of the electric currents involves the components of GmJ which can be written in the form GmJx = C,,0 G0eKxx sin kmz cos ky cos k,,' sin k,y' m=0 n=O oo co GmJyx = E Ej Cmn QeKyx cos kmt sin kny cos km,' sin kny' m=O n=O GmJxy = E E Cnn QeKxy sin kx,, cos k,y sin k,r' cos,ky' m=O n=O GmJyy = E E C,, nGeKyy cos kmz sin k,,ysil k,x' cos k y' (3.38) m=O n=O Notice that the coefficients are the same as for the Cel< case. It is not difficult to show that upon application of Galerkin's method, (K.mJETj) = (JiCeKw i) (.339) where the double inner product notation is defined by (at,El c,bj~) = | |c*bdS'dS (3.40) This implies, because of the signs of Equations (1.58-1.59), that the submatrix associated with the electric current contribution to the MFIE is the negative of the submatrix for the magnetic current contribution to the EFIE (diagonally opposite for the order given). This observation reduces the computational effort required since only one of these submatrices

needs to be calculated to fill their respective positions in the matrix. However, for the radiating slot problem when we use the Maxwellian transverse distribution. evaluation of the Sommerfeld integral has been accomplished through point matching which does not produce this symmetry. HIence, in that particular case, we cannot take advantage of this property. It can also be shown tlhat the electric current EFIE terms form a submatrix which is diagonally symmetric as are the terms in the diagonal submatrix representing the MFIE contribution of the magnetic currents. Therefore, these also can be formed by calculating only about half of the terms, however, as will be seen later, taking advantage of other matheniatical relations for tiese terms produces far more significant improvenients in the fill time for these submatrices. Cavity MFIE - Magnetic Currents K The cavity magnetic current hN[FIE terms are associated with a Green's function which can be expressed as 00 00 GC nl(xx = Z Z C,mn, mlixx sin k,z cos kny sin k,z' cos kny' m=O n=O 00 00 G mKyx = E E Cm,n mKyx cos km sin k,nysin km, cos kny' m=O n=O G mKxy = Cmn QmKxysin km: cos kny cos kmz'sin kny' m=O n=O GmKyy = E E Cmn, mKyyCOSk,,r sin knycoskmz'sin kny' (3.41) m=O n=o where GmKxx = Z[kZoo(i,C) + \ k Zoo(,im)] (3.42) GmKyy = n Z(~~) ( ki ) ~~(/m) (3.43) 5mmKyx = mlmKxy = -kmkn [Zoo(C) - (k) Zoo(fim)] (3.44)

60 The Z00 function is defined as [uiu sin kz(z - c) + j cos k:(z - c)] [tL sin kz(z' - d) + j cos k.(z' - d)] 00 [L sin k (z - k d) + cos k,(z - d)ii kz(Z'-c)+ jcos k(' - c)] } (~u - iL) cos k,(c - d)- j(MUL - 1) sin kz(c - d) for zz' (3.43) Slot MFIE - Magnetic Currents K For the slot, the transverse components of GmK for a homogeneously filled cavity are needed which can be written in the forms: G mKxx = i I Cmn Q,mKxx sin km, cos k,y sin k,Z'cos k,y' m=O n=O ZC 00 CmKyx = ~ ~ Cmn 5mKyx cos kmZ sin knysin km,'cos ky' m=O n=O GmKxy = Z E Cmn mKxy sinkmxcoskny cos kz'sin kny' m=O n=O GmKyy = E E Cmn 5mKyy cos k,, sin knY cos kmx'sin kny' (3.46) m=O n=O where cos k,(z' - d) k2 k2 5mKxx = sikk2(c-) k+ "+ mKyy sink,(c-d) k2+ sin k:,(c - d) m kj cos k, (z' - d) k~ (a.4 ) QmKyx = = 5mKxy = kkmkn sk ( d)[9 ( ) sin k,(c - d) (3-47) These expression are valid for currents on one end of the slot coupling to field points on the opposite end when t' = c. For currents coupling to the same end, z' is set equal to d. Half-Space MFIE - Magnetic Currents R Although the Green's function for the half-space was derived in Chapter II, the numerical treatment is quite involved and will not be detailed here. It consist of a real axis

61 Gaussian Quadrature schieme witli siingularity extraction of tlhe branch point and surface wave poles and asymptotic evaluation of large arguments. The methodology has been outlined by Katehi and Alexopoulos in 1361]. The numerical implementations used to evaluate the half-space admittance elements of the matrix were provided by iatehi for the Maxwellian transverse distributions and by Harokopus [31] for the rooftop functions. 3.4 Identification and Reduction of the Integrands The elements of each of the submatrices of Equation (3.17) involve double surface integrals as shown earlier. The inner surface iitegrrations are over'source' regions defined by the current expansion basis functions. The outer surface integrals result from the application of the weighting functions and cover tle'observation' or'field' regions of the problem where the boundary conditions represented by the integral equations (section 1.4) are being enforced. Using the condensed notation we can now write expressions for the impedance elements in a general form which will identify the integrations to be performed for each term. To illustrate, only the EFIE expansions for the current on the strip will be presented. The MFIE for the strip and slot currents are handled in an exactly the same maiiinner. The EFIE:electric current terms can be written as follows: Z', = E E>J f(V)(V)Jj, G (f, )' l(v)(vj) dS, dS, zA ll = E ( V )*("V)I J SGc"2') 4(vj),(v,) dS, dS, Z = GE E2|j (vi)(vJJG(2 V) (vj)() dS, dS (3.4 As before, the superscripts i and j represent the it "field" subsection and the jiL'source'

62 subsection. The Green's function terms can be transformed using the relations i = vcosq~- /sin0o 13 4% = vsin + cos< (3.50) The transverse Green's function, GeJ = Gejxxii + Gejyxy.i + GeJxy:P + GeJyyPP (3.51) can then be written for the strip-fixed coordinate system as GeJ = G evvf{, + G eJlvivl + G eJvuvO,' + G eJvv,vtl (3.52) where G eJvv = GeJxx cos2 4 + ( GCeyx + G eJxy) sin 0 cos q + G eJyy sin 2 G eJvv = -( GeJxx - GeJyy) sin 0 cos b + GeJyx cos2' - GeJxy sin2 G eJvv = -( GeJxx - GeJyy) sin cos 4 - GeJyx sin2 + GeJxy cos2 G eJvv = Gejxxsin2- ( GeJyx + GeJxy)sin cos + GeJyy COS2 4 (3.53) When the coordinate systems are mixed, such as the case of the magnetic slot current contribution to the EFIE, the Green's function terms are also mixed. The relations for the slot-fixed coordinate system unit vectors are i = (cose- sine (3.54) = (sinO+~cose (3.55) Substituting these into the posterior positions of Equation (3.51) and (3.49,3.50) into the anterior positions, the Green's function terms for this case become G eKv( = GeJx COs bcos + GeJyxsin4 cos + CGeJxycos sin + GeJyysil o Qlr t

63 G eKvu = - G eJxx cos sin 8- GeJyx sin d sin G + C; eJxs cos 0 cos + C;,J!. sin o cos 6 G eC ( = - GeJxx sin6cos + Gejvx cos cos - GeJxv sin ( sin 8 + C; eJvy coS ) sill 0 G eKv, = Gejxx sin in sin- GeJyx cos sin 9 - GeJxy sin 0 cos + CeJ. os O cCOs (3.56) Returning to the EFIE:electric current case, we can now write the terms of tlhe associated submatrix terms as (omitting the C,, constant) 00oo 00 Zel = Z seJx- CoS2 ~eo(Vl V"l)Ieo(Vj I'J) + QeJyx sin fdcoso~e(t"',,' )[eo(t,j v, ) m=0 n=O + eJxy sin 4 cos I,,o(vu,, )loe( vj, v) + 5eJyy sin2 l!oe(',. vL )I,o( lJ, Lj ) Zj' = C -Z eJxxsin f cos lIeo(Vi,v:)Ieo(tv,)LJ) + GeJyx COs2 [e(v,.'i, )[eo,,( v, j ) m=O n=O - eJxy sin2 GI(ejxy, vi )oe(v j,) + 5e sinsin 0 cos O,oe(V,. L', ) I,loe( v. ) 00 00 Z2 j j l —eJxxSin cos veo(v,I, )eo(v,vj ) - GeJyxsin2[oe(V,')[eo(Vj,t I'Jj) mrn=O n=O + eJxy cos2 leo(vi, Vi)Ioe(j, vj) + GeJyy sin Ocosf[oe(v,, -l, )loe( vi, t'j) 12= Z Ge QJxx Sin2,co(Vs,v,)Ivo(L',V.,)I - ve-yxSilneyOsinoe(VI, V)Ieo(VJ,Ij) m=O n=O - GeJxy sin cos e(v, vi)) + yy oe(,j) + eyy cos2 loe(,,'i) j, vj) (3.57) Now it can be seen that the only terms involved in the integration are of the form I0(Qq6cq, ) c=skz o sin ky j ](aQ)q(3q) dSq (3.58) d sink-, z cos k.n/ where (aq,, q) could be either (v., vq) or (vq, vq) with dSq = dt,dv for the strip or (aq,B q) would be replaced by ((q,tq) or (q,C,q) with dSq = d(df for tile slots.

3.5 Integration We will ultimately assume, as is commonlv done, that the strip and slot are sufficiently narrow so that tile longitudinal components of current are much greater in magnitude than the transverse components. The latter can then be neglected which, as will shortly become apparent, greatly simplifies the book-keeping required to keep track of various Green's function and current components, coordinate transformations, cross-coupled terms, etc. The assumption is further justified in that, at this point, there is no known advantage or requirement for the microstrip or slot structure to be more complex. For the moment, however, the complete expansion will be retained so that the numerical model can later be extended based on these expansions, by evaluating the additional terms of the matrix. Now the Io(al,,/q) fuinction is still a mixed coordinate system function, therefore, to perform the integration the function must be transformed to a common system. The simplest approach is to transform the cavity-fixed coordinate system functions into the stripfixed system by solving Equations (3.7-3.8) for x and y and substitute into Equation (3.58) giving z = vcos - vsin + xi (3.59) y = vsin-+vcoso+y (3.60) Equation (3.58) then becomes I~o< (a Ip ) cos[k(vcos X - v sin 4 + x.)] sin[k,,(vsin X + v sill 4 + y)] \ Js I sin(km(v cos v- vsin46+ zo)l cos(kn(v sin 4 + vsin X+ yo)) I *4(aq)* p(O.q) dvdv (3.61) By introducing the notation, ki = kmcos ~iftsin4) (3.62) 1 - = kmsinf)ikncs4) (3.63)

6.5 Vo = k,mo. L, = k,yo we can reduce these integrals to the form c(aq,/qi) = 2 s1 I [sin(k+v + vo) cos(k v- vL ) - cos(kv + vo) sin(k-v - L' T sin(k- v+ vo)cos(k+ v + vo) cos(kv + o)sin(k+ v vo) ~(aq)(0,3q) dvdv (3.66) It is now clear that there are only two integral forms which must be evaluated: Jsin(koacr + ao) cos(k03 +,30)4(a)T(P) da d3 j/ cos(koa + ao) sin(ko3 + 3)(a)q( /) da d (3.67) The integrals involving 4(a) are expanded and evaluated in straightforward fashion to give [ csin (ka si) (a) dCOs (akb ~(kqao)] sinc [(k + kb)- 2 siiic [(k - kb )ij (3.68) where sinc(z) = sin(z)/z. For the case where I(,O) is the Maxwellian distribution, 1~ - mkO ~|ao(W Oo)] ( = lf'| [ osin)d(kO i 0~)] / a 2(3.69) Co[ Ssin + P, (3)da= VIt) Cos1Using the substitution = sin Y, we have b= /J/[ (klO sin7oy )] dy -r,./2 cos _ os~o)~0, s/2, COS( sin sn i, c Cos ] c c,/os(3' )-12[ (k10 sin 7)] d) (ksin)] d (3.70) This form can be reduced to I = sin (i/30)] cos(klpsin'7)d = [c (:P:o)] Jo(kI0 (3 1)

66 For the rooftop functions, the transverse weighting is 1/21B so that =J = sin (kj3 4) 30) H(')d:3 sin (k323 30) d, IC (~23) sinc(kb1) (3.72) With these results, the integration of Equation (3.G6) is fully specified and we can proceed to evaluate the terms of the matrix. 3.6 Numerical Evaluation Considerations Before proceeding to the applications, somne comments on thle numerical implementation should be made. At this point we are in a position to go ahead and program the previous expressions to evaluate the matrix elements as they stand, however, just a brute force approach, without some consideration of the algorithms to be used, would undoubtedly result in a very inefficient program whiicli takes muchl longer to run than need be. In this section, some'common sense' features will be pointed out in addition to some matheniatical identities which can be used to significantly improve the convergence rate of the summations involved. 3.6.1 Precomputation By writing out the complete expression to be evaluated for the self-impedance on th4 strips, the main points can be illustrated which also apply to other elements of the matrix Let us assume that the strip is oriented along the i axis in the cavity with a width of 1T and is centered at the point (z, yo). A typical element of the [Z11J can then be written a m= z j I (z) )(Y/) G(11) (x')*(y') dz dy dz'dy' = Z Cmn eJxxleo(i, Yi)eo(',. Y:i) (3.7: m n

67 Assuming tile Maxwellian transverse dependence, the'eo terms are evaluated as In;t1 previous section and Z,, becomes (I Z) Cm k2,4 ZI) ) C E sin2 kbl 5eJxx cos(k,(x, + x,,)] cos[k, (I + I )] sin kb1, m n sin2 knyoJ2(knly)sinc2[(km + kb)lz]sinc2[(km - kb)lr]( 3.74) The most basic rule to optimize the speed of the computations is to perform any operation or function evaluation as few times as possible. Immediately we recognize then, that thle conistants sliould be factored out and the outer loop factors shiould be removed from the iniller loop: 2b r -l 4fl E (2 cosk ) m( + xZO)] cos[km( z +'o)) ab sin2 kbl n 2 C0k( sinc2[(k, + kb)l1]sinc2[(km - k6)l] * {Z ejxx sin2 kn,YoJ2 (knl}) (3.75) Now in this expression, although evaluation of the sine and Bessel function J.' in each cycle of the loops is implied, in practice these are computed and multiplied external to both loops and stored as a vector dimensioned to include the maximum value of'n' to be evaluated. Thus, these functions are evaluated only once per value of'n'. Similarly, the (2 - 6,)/2 factor ( = 1 for m $ O ) appears only in the'm' loop, since for n = O there is no contribution (note also that the conditional statement whiclh tests for m = O can be eliminated by calculating the m = 0 terms separately). 3.6.2 Transformations Finally it is noted that the'm' loop implies m x N evaluations of the cosine terms and with the inner n loop implies m x n x N2 products of these terms and the remaining terms where N is the number of basis functions on the line. This can be dramatically reduced

by making use of the relation cos A cos B = -[cos(A + B) + cos(A - B)] 3.76, so that our expression becomes - I 2f pk214 (2 - 6m) z' =2k;bt E (2 (cos(k,(z, + x: + 2z:o)] + cos[k,,(,: - Iv )])'J ab sin' kbl,, 2'sinc2[(km + kb)l]jsinc2[(km - kb)l] {Z eJxx sin2 kryo.IJ(kn,~)} (3.77) We now need at most m x (3N -1) cosine evaluations, on the order of m x n x N products and VN2 simple additions which greatly improves the computation timne, especially as N increases. The sum resulting from the application of Equation (3.76) is performed after the m and n summations are complete. This is the key to the speedulp since a factor of N is removed from the number of product evaluations. A factor of N2 additions are add(led, however, this is inconsequential since the additions are simple (two complex numbers) whereas the previously required product evaluations also involve the calculation of other non-trivial coefficients. If the matrix elements are to be stored in data files, storage of tile cosine sum and difference terms also greatly reduces the file size - now on the order of (31V - 1) rather than N2. This technique also applies to the slot self-impedance terms. For the coupling terms (slot-to-strip, etc.), the same principles are used except the sum and difference scheme no longer applies due to mixing of km- and k, wavenumbers. However, in these cases, due to the separation of the structures in the i direction, sine or cosine factors in the denominators become hyperbolic for large values of m and n and improve the convergence rate so tlla the upper limits of the summations can be reduced. Other factors can be considered, however, the above points are thought to contribute the most significant improvements in program efficiency with minimal effort. Of course, there are other issues which have not been addressed since they often depend on the avail

69 able facilities; memory limits for example. There are also schemes available to inmprovo convergence thirougli auxiliary series transformations, hlowever, these are often onl! possible at the sacrifice of generality. For example, for uniformly filled cavities. e.g.. stripline problems, tecliniques such as those found in [13, Appendix A.6] can be applied but then multi-layered cases would require separate treatment. Admittedly, thle full extent of these possibilities have not been adequately explored and may offer further improvements in program efficiency. Rather, the emphasis has been focused on phenomenological exploration of the applications in thle discussion to follow. 3.6.3 Convergence, Algorithms and Run-Time As with othler eigenfunction expansion methods, consideration of the convergence of the modal summations is an important process in establishing reliable results. For these types of solutions it would be desirable to analytically examine the expressions involved and derive formulas for acceptable upper limits of the summations. Ideally these formulas would be provided for each type of Green's function and would be functions of all the relevant geometric and electrical parameters and the desired accuracy. To develop such a system, however, is a major undertaking in itself and would really only be worthwhile after a more thorough investigation of possible series transformations alluded to above. In addition, convergence behavior generally varies depending on which output variable is sought, further complicating the situation. Nevertheless, it was necessary to investigate some aspects of this question in order to produce reasonably efficient programs, the intention here then being to pass along some of the information gained to those who may extend the scope of this work. Also it should be recognized that although it has become common practice to discuss convergence by showing the behavior of particular parameters for the structure at hand as a function of the number

70 1~0.0,...........I....I....i.... 10.0 8.0 -03 —-. Effeaive Rcl,aive Dielcnc Con.st 6.0 _ 6-0 ChQarac.en'stic Impedaace 4.0 \ ---- Tnnsverse Tangential H field on Shield Below Strip 2.0 -8.0 ^ # -2l.0 0 25 75 00 25 50 values at 1000 modes were ZO = 49.63Q, r,el.. = 6.90, Hx = 12.45A/A. of modes used in their calculation, this information is really of limited value since when the technique is applied to a different structure, there is usually no guarantee of similar results. The plots given here are therefore provided only to illustrate some intuitive points and to give the reader an'order of magnitude' feeling for the required range of upper limits. Unless some analytical guidelines are developed and become available for these problems similar numerical experiments must be performed for each new application. To illustrate, let us first look at the convergence rates for several transmission line characteristics. The theory behind the calculations is presented in Chapter IV. Figure 3.4 shows the convergence behavior for three key transmission line parameters of a shielded microstrip line. As can be seen, the rate of convergence depends on which characteristic is to be computed. We re-emphasize that that these features may vary as a function of the

71I geometric and electrical specifications. It shlould also be mentioned that tiis appiication does not present any practical difficulties since not only does the convergence appear to b,o quite rapid but tlhese parameters are computed by a two-dimensional formulation involving only a single summation. The efficiency of evaluating the summation for this problem is not a significant issue given todav's desktop computer capabilities. For example, the data in the figure were efficiently computed on an Apollo DN2500 worlkstation. This brings us to the consideration of the two dimensional summations required in the analysis of three dimensional problems of the type discussed in Chapters V-VII. The required number of terms in the double summations depends on their behavior in the mn-plane, where m and n are thle two parameters of sumnlation. Thle real problem is in predicting or anticipating this behlavior a-priori so that the corresponding limits can be set. Some progress in this direction call be achieved by recognizing that these summations are similar in many respects to thile Fourier series. By drawing on our understanding of this topic, we can gain some intuitive understanding of how the modal summations behave. For example, it can easily be shown by numerical experiments that the closer two elements are in spatial coordinates, the greater will be the extent of the mode spectrum as in Fourier analysis. In addition, there may be no sign changes for the self coupling terms so that we can immediately conclude that these terms will display the slowest convergence. Thus, we could potentially monitor only the self coupling and nearby terms to determine whlether the summations have converged or not. There is also a potential savings in time, if we can monitor the convergence of the series as a function of one direction (referred to as an'eigendirection'), the direction of m say, while the other (n) is constant. Thereby we can potentially eliminate significant fractions of individual rows or even entire rows, depending'The Apollo DN2500 us quite slow relative to most other engineering workstations, perhaps comparable in speed to standard Intel 80386 baed personal computers. The above data was computed at a rale of approximately one minute cpu time per 1000 modes. The relation is approximately linear since there is only a single summation.

72 on the beliavior of the functions involved. While these ideas liave an attractive generality to tlhem in tlhat we mighlt be able to develop an algorithm for monitoring the convergence of the sum, making the convergence question invisible to the user, there are serious difficulties when we try to put it into practice. Most significant of the pitfalls involved include the fact that the final result is not known a-priori so it is quite difficult to establish a criteria for convergence. To illustrate. consider what happens if a'percent change' criterion is used. Suppose some of the terms of series at the early stages of summation contribute very large values whlich are later canceled by similar terms of the opposite sign. At the early stages then, the sum will be large and a fixed percentage of the current sum may be quite large compared to the final result. Thus, truncating the local contributions to the series based on the current percent change may prevent important terms from making their contributions which in the end leaves a significant error. Another problem is that, as with Fourier analysis, the mode spectrum may have extended gaps along the eigendirections so that a summation in that direction may appear to be converged when in fact there are addition terms farther along which are needed. Some time was invested in pursuing an algorithm which takes these factors into account based on observations of the spectra of various cases. In the end, however, it was found that while some progress could be made by instituting various monitoring schemes, the final conclusion was that the overhead required for keeping track of the progress of thle sum and the periodic testing of sum's status, more or less offsets the gains made in reducing the number of actual terms added to produce the result. However, one scheme that emerged does bear mention here since it is straightforward. It should be recognized that the cancellation of terms generally occurs along both eigendirections and also the largest terms occur near the origin. Then, one can readily accumulate the bulk of the sum by making sure

to sum thle lowv index terms near thle origin at tile onset, tlhus providing a good estimate of the final result and alleviating the problem of not having a good estimate of thle final result in advance. A very simple algorithm accomplishes this scheme by simultaneouslv incrementing either m or n while decrementing the other and successively moving away from the origin, thereby summing in a direction normal to the diagonal of thle nin-plane. However, because of the tradeoff mentioned, the final versions did not use an) of the developed schemes but simply scanned the mn-plane in a straightforward, raster-like fashion with terminating conditions set by experience. Generally, to avoid any question of convergence while other investigations were underway and because sufficient processing power was available, far more terms than necessary were used anyway, typically on the order of 1000-1500 modes in the eigendircctioii associated with the directions of the lines or slots, and half as many in the other direction. For the cases studied here, not nearly as many modes in the one eigendirection are needed as in the other. This is due to the fact that in the particular cases studied for comparison to experiments, the strips and slots were always parallel to the side walls of the cavities so that one coordinate describing the position of the basis functions is constant. This tends to cause the spectral variations in the corresponding eigenvalue to be similar for all basis functions, although the variation of the other coordinate prevents this from being strictly true. If, for example, the y coordinate is constant for all basis functions and n is the associated eigenvalue, the spectra in n at a fixed value of m will be identical except for a constant scale factor which depends on x. Truncation error in n then tends to get averaged out by the variation in z. To illustrate, let the aspect ratio in this example be defined by NIM, i.e., the denominator is the maximum eigenvalue corresponding to the coordinate which varies and N is the maximum of the other eigenvalue (N < M). Figures 3.5 and 3.6 demonstrate a typical convergence experiment in which the aspect ratio has been fixed

-~. 6.0 u t~~ [_o~ ~~~ 0Aspea RaLo =.5 5.o 4.0 3.0 2.0 1.0 L 0.0. 0 250 500 750 1000 1250 1500 Number of Modes (mn) Figure 3.5: Convergence behavior for peak normalized resistance versus maximum mode number. at 0.5. The quantities examined are key parameters to be introduced in Chapter VI. As always, othler parameters may converge at different rates and the rates may vary as geometric or electrical parameters change. The key point here is that at least for this case, when the maximum m value is set to M = 600, the changes which occur as N/M is reduced from 1.0 to 0.25 are less than 0.1% for the resistance and 0.03% for the resonant length. Thus, far fewer modes are needed in the n direction. For subsectional basis functions, the spatial sampling rate also requires convergence criteria. Since the quantities we are dealing with, e.g., impedance, are typically highly sensitive to the behavior of the near-field and also often depend only on the fields in a small region, the required spatial sampling rate is generally substantially higher than what is required for far-field type problems (or non-uniform sampling is needed which is more complex to implement). Again, the actual requirements depend on the circumstances

0.000.... i I -.100.= -.200 f._ t Aspec RaLio.5 LLU -.300=....!. A I t 0 250 500 750 1000 1250 1500 Number of Modes (mn) Figure 3.6: Convergence of resonant slot length as a function of the number of modes. and techniques used so it is difficult to provide general statements on these requirements. Twenty samples per material wavelength is often used as a'rule of thumb' for far-field problems which we generally increase to the 30-40 samples/wavelength range for sampling on strips or non-resonant slots. For the resonant slots, we often increase the sampling to 100-125 samples per wavelength making sure to overestimate the resonant length of the slot by a significant amount. This practice is not driven so much by convergence criterion as it is by practical matters. It takes far longer to generate the matrix elements than it does to invert and process the cases studied here. Changes in length or the relative positions of slots and lines for example, can easily be accomplished by loading a matrix with longer than needed lengths and scanning the behavior of the structure as a function of lengths or distances by successively removing the appropriate rows and columns; inverting and solving for the appropriate parameters at each stage. Thus, the matrix elements

76 need to be generated only once for a given frequency, allowing a wide variety of geometric variations to be studied as long as the cavity dimensions are not changed. We also use this technique to provide greater resolution of lengths and relative distances without requiring a regeneration of the matrix. The upper limit of the spatial sampling rate when the elements of the matrix are computed with double precision seems to be on the order of about 250 samples per material wavelength. This number, attributable to round-off and truncation error, is machine and algorithm dependent which brings us to the final point on this subject. For problems of these types, it is common practice to quote run Limes typical for certain machines, often not very well identified. It is common knowledge that machines vary widely in their ability to execute codes, especially from machines of one type to another, but even for the same type of machine with different hardware configurations. In addition, the information provided can quickly become dated and irrelevant since available computing facilities are changing rapidly. Not only this, but also the execution time of the same analysis implemented with different algorithms and different degrees of generality can vary widely (easily on the order of a 10 to 1 variation, depending on both the abilities and knowledge of the programmer as well as the time spent in optimizing the algorithms for speed). Undoubtedly, there are techniques which can be used, in addition to those discussed above, to improve the run time of codes developed for this analysis. The real issue comes down to a tradeoff between time spent on optimizing codes versus producing and examining results. Of course, the outcome of this tradeoff depends on available resources, the objective in producing the codes, and their intended end use. Suffice it to say that the key element in the current approach is the generation of the matrix, which for a typical three dimensional structure at a single frequency, can be generated in 45-60 minutes or less on a - 25 million instructions per second machine such as the IBM RS6000/320. This amount of processing time has been sufficient for our needs.

CHAPTER IV ANALYSIS OF TWO-DIMENSIONAL STRUCTURES The treatment of mtiltiple layers can be illustrated in greater detail by presenting the methods used to analyze structures which are uniform in one dimension. Complex threedimensional problems are ofteii treated initially in this manner by analyzing tlieir more fundamental elements - two dimensional transmission lines - with lumped elements added to represent discontinuities. The total problem can then be treated by network analvsis if the basic properties of the individual structures are known. The objective here is to show how these fundamental properties can be obtained, for instance, the propagation constants and characteristic impedances of transmission lines. These quantities are directly tied to the solutions for the fields in the structure which thus becomes the main objective of this chapter, that is, to demonstrate the procedure for matching field components through multiple substrate and superstrate layers as applied to two-dimensional problems. The presentation also serves to illustrate how the technique can be applied to more general three-dimensional problems since the procedure is the same. Some examples of field solutions are given to demonstrate the utility of the approacih. The work in this chapter is also needed in later chapters which deal with the microstrip-fed slot antenna elements. The scattering of waveguide modes by vertical wires is also studied in this chapter. Tills problem reduces to two dimensions for the homogeneously-filled waveguide case wIidi,

is relevant to the experimental work on stripline to be discussed later. The problem is unique in this work in that the currents are normal to directions of the other currents discussed. The treatment of wires inserted through multiple layers cannot be reduced to a two-dimensional problem and will not be discussed although, as will be seen, the solution to this problem would be needed to complete the general analysis of waveguides and especially cavities formed by inserting wires through the ground planes of multi-layered parallel plate waveguides. It is also noted that even for the homogeneously-filled case, thie problem of vertical posts or wires has not been extensively treated in the literature. especially for parallel plate structures, even though they are often used in practice for the suppression of unwanted higher-order modes. 4.1 Application to General Multi-layered Shielded Microstrip Structures The objective of this section is to show how the fields in the various levels can be calcu lated from a known form of current density on a single microstrip line in a straighltforward way. Also, although to this point we have considered multiple components of current, tlle applications to be discussed will be restricted to narrow strips with one current component to minimize the complexity of the presentation. There is no restriction on the placement of the current or the number of strips which can be used, however, a simple case hlere will better serve to outline the method. The procedure for the treatment of more complex multi-layer coupled strips can be found in [83], which serves as an example of the use of potential theory with impedance boundary conditions to generalize multiple layers in a way similar to what has been done here. This reference also addresses the modelling of strip conductor loss. In additon, we compare to a much earlier work on this type of analysis by Yamashita [90], whose method of non-uniform discretization would be appropriate for wider strips where the form of the current density cannot be assumed.

79 In order to calculate tile characteristic impedance, tile definition Z = P/II is used where P is the time average of power propagating along tile guide. Thus. if we set / =. the cliaracteristic impedance is simply Z = P which can be computed analytically by integration of the average Poynting vector on the cross section of the waveguide. Also, in some cases, the reaction of the fields (R) on the waveguide cross section is needed which can be computed ill the same way. For our purposes, let us assume a geometry sucli as shown in Figure 4.1 with a longiz................................. X.............. X0 x=a Figure 4.1: A shielded stripline, uniform in the y direction with multi-layered substrate and superstrate. tudinal current component on a single narrow strip of the form 26(z' - d) " I' - zo < W/2 (4.1) J(Z', y, z0') (4.1),rW /- 4 (z'-)2 i.e., a Maxwellian transverse variation of the current density whlich satisfies the edge conditions on an infinitesimally thin narrow strip. The propagation constant k, represents the set of characteristic complex phase constants associated with shielded microstrip modes.

8o These complex constants are the eigenvalues of the equation Et = -jp JJ x GeJ -J dV' = n x Z, J ( 4.2 which enforces the appropriate boundary conditions on the surface of the microstrip line [83]. A particular value for k9 must be found first, before the fields of a particular mode can be calculated. Using Equation (4.1) in (4.2); the y component of GeJ as derived for layered rectangular waveguide in Chlapter II; and the integral representation of the delta function, / e_j(,_tk)Y dy' = 27r6(k9 - k,) (4.3) 00 we find that the y component of the electric field can be written as +00 00 (2 6-m)6(kk- ky) E I+c moE (2 )(k - kv) sin kxz sin kxzoJo(kW/I2).frk2 [ tJcL[(7eU cos kz(C - z) + j sin kz(c - z)] (leu - e(L) cos kz(c- d)- j(eUIriL - 1) sin k(c- d) k22,L[m v COs k(C - z(c) + jsin k,(c - z)l (4.4+ + I (fmnU - imL) COs k(C - d) - j(fim mL - 1 ) sin kz(c - d) ( where the Fourier integral can be eliminated using the sifting property of the 6 function. Then, using Galerkin's procedure to enforce the boundary condition given by Equation (4.2), we can write = a - W (2 -,) sin2 oo2( k a k3(k2 + k2) 2 [ TL(TU cos kt(c - d) + j sin k,(c- d)J { L([c4- cL) coS k,( - d) - j( L 1sink(c- d) +k2 [i -,m,L[)c osk,( c'- d')+ j( sink,(c - d)s - d) (45) k, (4mU - 4YnL ) cos k,(c - d) j(4mU4nL - 1) sln k,(c-d) We now numerically search, using Muller's method for example [60, p.2623, to find the values of kg which satisfy Equation (4.2) and thus correspond to microstrip modes. With the propagation constants known, the fields in the i'h layer can be easily found. By inspection of Equations (2.80) and (2.85), the fields can be written in a general form

81 in terms of the homogeneous solutions to Equations (1.10) and (1.12). Thus, the tiolds n the ith layer can be written as 0 j(2 - 6,m)ki Hj - Z 27rak,(k2 + k2) imu Moo(k, ks(z - d)] + jMMoe[kr, kz(z -c) B im HLMoo[kz, kz(z - d)] + jMoe[kr k,(z - d)] ( [7cUNee(kzs kz(z - c)] - jNeo[k;r: kz(z - C)l] + A, (4.6) [feLz Nee[kz, kt(z - d)] - jNeo[kz, k,(z - d)] J and -, j(2 - 6m) EJ - Z 2rakz(k2+ k2) ( u [eUMee[k k(kz - c)] - jI40[k., kz(z - c)]] L 4 Mee[k k:(z - d)] - j4o[k., k,(z - d)J] (-u [71mU [k k,(z- c) + jN.[k,, ks(z-c)]] + Bi (4.7) + [B mLNoo[kz kz( - d)] + j,>[k., k,(z - d)] which are valid for all layers except the source layer. Some of the leading constants are preserved for convenience in later notation. 4.1.1 LSE Modes Expanding the VWFs we then find that the LSE mode fields in each layer can be written in the form 2wA1 ELSE = A [akki(x2; k;2) [ |zjk-ky cos kxz sin k.x' - ijk- sin k,z sin kz'l cos 1,(c - d) cos k,i(c - z) [f,eu + j tan k(ci - z) (i - 0) cos k(c - d) cos k,i(z - dz) [i/CLi - jtan kri(z - d,)] (i > 0)

82 HLSE [a(, k: k sin k z sin k z' + y. yjk k, cos ki si n k z { cos kz,(c, - z) [CUI, tan kz,(c, - Z) -;1 O0) cos ki,(c, - d ) | cosk, (z - d-) [r7jLi tankr1(z - d ) +] ( 0) +A1 [.] k. cos k.z sin k.z'] 1 J cos kt1(c, - z) [fi,u, + jtank, (c, - z)j (z < 0) cosk~,(c1 d1) cos k,,(z - d)(r os,L, - tan k,,(z - d,)j (z > 0) (4.9) where i = 0 for the layer containin' the strip and Ao = (4.10) (Jeuo -.eLo) cos k,(co - do) - ]( ii,uo~ - 1) sin k1,(co - do) By matching the tangential components of the field at the interfaces above and below the strip layer the remaining coupling coefficients are found to be A =I AO kz [euo + j i tan kzo(co - do) 1 kzo IzeLl - j tan k,l(c, - dl) f Ai-lk-t sIeL(I -l)(i > 1) A _- k,(i-l) cos k,(j-l)(ci-I -di-jl)[,.i -j tan k,i(ci - d,)] (A < O) kz(i+1) cos k,(i+l)(ci+l - d+l)[ieUvi + j tan k,,(c, - d,)1 (4.12) For computation of power (P), the needed component of Poynting vector is Y 2 E x W while for reaction (R) we need y- E(ky) x H(-k.). For the case of LSE modes, power density and reaction are given by P =E~H - _ 2wp [ Aiksk; 1[A!kZ l cos2 k,x [f sin kzx'12 = r. 2= 1k -(k2 k2)+ ) k; j Icos ki(ci - d,)i2 l ui cos k,i(c, - z) + j sin k,j(ci - z)12 (i < 0) 1J,U cos kl,(z - d) - sin k,i(z - d )l2 (1 > O) (4.13)

83 R Ep [kA,1H kk + k A2 ) k] jk ] cos2 kz [f sin k:,'!2 R EA(k,)H,(-k,) = zT kz,(k2+ k kk, COS2 k.,(c, - d,) [7jue; cos k~,(ci - z) + j sin kz,(c, - z)]2 (/ i < 0).<~~~~~~~~ ~~~(4.14) I [OLi cos kzi(z - d) - j sin k~,(z - d,)]2 (i > O) 4.1.2 LSM Modes Similarly, the LSM mode fields can be written as ELSM = -Bi ij -kzk LZ( < 0n ELSM = -B ak2(k2 + k 3 Ykk9 cos kx sin + 2 sin sin z' { cos k,i(c - z) [iimi + j tan kz,(c - z)l (i 0) cos k..i (ci - di) cos k,i(z - di) [rlmL - j tan k(z - di)] (i > 0) ~B, [2ikM] [ijjk, sin k:z sin k,x'] t cos k,i(ci - z) [Omui tan k,i(ci - z) - jJ (i < 0) cos kzi(c. - dj) |cos kzi(z - d) [mLi tan kzi(z - d) + j] (i > 0) (4.15) HLSM Bi [)] [=k2sin k:x sin k(kx' - )jk:kv cos k,z sin kx'] 1cok2(c di){ cos k,(ci - z) [(iui tank,i(ci - z) - j] (i < 0) cos k,i(z - di) (fimj ta k,i(z - di) + ij (i > 0) (4.16) where the coefficients are defined by Bo = -~MLO (4.171m U = (OmO - m) cos ks(co-do) - j(?mUO77mLO - 1) sink,(co -do) (4.17) B1 = B [ LUo+tank(4.18) k Bi lk(+)kmU( kZ(i+l) cos k,(,+l)(c,+l - d,+l)[rii,u + j tan kz,(c, - d,)] ( 0) (4.19)

84 To calculate the field at a particular point in the ith layer, we then calculate only the coupling coefficients (Ao... A,) and (Bo... Bi) while evaluating Equations (4.S.4. )! and For the LSM modes the required terms for calculating power density and reaction are given by, 2w [Biky ][ B (k;)2 1 sin2 k,z [f sin kzx']2 p =-EH = -a2L kJ (k2 + k2)- I cos k,i(c, - di)12 ~ { ImUi sin k,i(ci - z) - j cos k(c z) (i < 0) I7?mLt sin k,i(z - di) + j cos kz(z - di)l2 (i > 0) (4.20) 4wY [jBik,] [ -jB,kv ] sin2 kzx [f sin kxz']2 R= E(k)H(-k) = - -a2 k? (k2 + k2) cos2 kz,(c, - d2) { [Im,ui sin k,i(ci - z) - j cos ki(ci- z)]2 (i < 0)'[,T,mLi sin k.i(z - di) + j cos ki(z - d)]2 (i > 0) (4.21) 4.1.3 Cross Terms There are also two sets of cross-terms from the ELSM X HLSE product. The power terms are 2wp Bikz,kki] Ak, cos2 k,z [f sin kzI]2 Pa2 k(k2 + k2)] k j Icos kz(c - di)2 [imUi cos kzi(ci - Z) + j sin k,i(ci - )] [Ui cos k,i(ci - ) + j sin k,i(ci - z)] (i < 0) [,mLi COS ki (z - di) - j sin k,i(z - di)] [(Li cos k,i (z - di) - j sin k,i(z - di)]' (i > 0) (4.22) P C= -EH -I 2wp rBiky r Ak,k2 1 sin2 k,z (f sin ks2'J2 2 kl (k2+k2)]- Icosk(ci - d,)l12 [ [,mu, sin k,;(ci - z) - icosk,i(ci - z)][ f.u sin k,(c, - z) - j cosk,(c, - z)]' (i o) [imL sin k,i(z- di) + j cos k,i(z - di)] [k,i sin k,i(z - di) + j cos k,,(z - di)]' (i > 0) (4.23)

85 For the reaction we have R -/ E H ( -k) = 4 jBikzkykzi] [jA:kzl cos2 kzx [f sin kzX']2: - E'(') a2(-k-~) [ k2 (k2 + k) JL k J cos2 kz,(c1 - d;) [7mUi COS kz,(c, - z) + sin k:i(c, - z)] [u; cos k(c - z) + j sin kzi(c1 - z)] (i < 0) (,mLi cos k,i(z - di) - jsin kzi(z - d)] [Li cos ki(z - di)-j sin kzi(z - d,)] (i > 0) (4.24) 4-L, [jB,ky,] -A,k2 1 sin2 k_z [f sin k:x']2 2 [ k2 (k2 + k2) cos2 kz(c - dc ) { [mU sin k, (c, - z) - j cos k,i(c - z)] [eU,i sin kzi(ci - z) - j cos k(c- z) (i 0) [imL, sin kz,(z - d) + j cos kz,(z - d)l] [itL sin kz,(z - di) + j cos kzi(z - d)] (i > 0) (4.25) 4.1.4 Integration of Power and Reaction terms The evaluation of total power can be done analytically by integrating the power density on the waveguide cross-section. Integration of the z dependence is trivial and has been indicated in the previous expressions. The z dependence appears in two forms, the first of which is [rh cos k(c - z) + j sin k(c - z)][r2 cos k(c - z) + j sin k(c - z)l',PM = dz 1 [7 cos k(z - d)- j sin k(z - d)]j2 cos k(z - d) - j sin k(z - d)' 77i4(7) -sin 2e(k)(c - d):F jR( - sin2 Re(k)(c -d) rhan1 +1 rll + rl sinh2 m()( - d) + 477 (k)sinh 2Zm(k)(c - d) 21+7sinh 2 m(k)(c-d) (4.26) 4Zam(k), Tam(k) where XRe(k) and!m(k) symbolize -the real and imaginary parts of k respectively. The second form is c [ rl sin k(c - z) - j cos k(c - z)][2r sin k(c - z) - j cos k(c - z)]' 1 IPN = dz Jd v, sin k(z - d) + j cos k(z - d)ll[2 sin k(z - d) + j cos k(z - d)]J = 14~(k) 1sin 2Re(k)(c- d) - j - sin2R(k)(c - d) +4R9e(k i-) 273e(k) k)(k) 4~m(k) 2ZSm(k) sinh Ztm(k)(c-d) (4.27)

For reaction we also have two types of integrations to perform. The first is c [ri7l cos k(c - z) + j sin k(c - z)]7[r2 cos k(c- ) + j sin k(c - z)] [r771 cos k(z - d) -j sin k(z - d)][Th cos k(z - d) - j sin k(z - d)] J = 7 l(c - b r/712 + sin 2k(c - b) + sin2 k(c - b) 4.28) 2 4k 2k Similarly, c ['l sin k(c- z)- j cos k(c - z)][ sin k(c - z) - j cos k(c - )]1 IRN = I d [r/7 sin k(z - d) + j cos k(z - d)][r72 sin k(z - d) + j cos k(z - d)] 11772 - (cd) -77T172+1 sin 2k(c - d)' + t2 sin2 k(c - d) (4.29) 2 4k 2k 4.1.5 Applications With these analytical results, the fields, power flow and reaction, characteristic impedance and propagation constants can be readily found for shielded strips with any combination of layered substrates and superstrates. An example is shown in Figure 4.2 where results of the present technique are compared to a commercial CAD package (Touchstone [20]) and experimental measurements provided by Dunleavy [19]. The relatively large error bars provided by Dunleavy for this case do not provide for any conclusion on the comparative accuracy of the full-wave approach since the agreement is excellent. The corresponding characteristic impedance is shown in Figure 4.3. We conclude that the accuracy of the full-wave implementation is excellent since the accuracy of Touchstone is well established and has been further verified by Dunleavy for this case. One can argue that the full-wave implementation with its greater complexity is not needed, however, most CAD packages, including Touchstone, are based on approximate formulas which, although they can be quickly evaluated, generally decrease in accuracy as frequency is increased. Also they deal almost exclusively with the dominant propagating

87. 7.40 7.20 - U 7.00.U. 6.80 u 6.60 > 6.40 2-D Full Wave Frequency (GHz) Figure 4.2: Effective relative dielectric constant for shielded microstrip on alumina (a = b = 250 mils, WV = 25 mils, e. = 9.7 ) compared to measurements [19] and Touchstone [20]. 55.0 - X!,, 54.0 Dual. 9C 53.0 N 2-D Full Wave 6..........._TbW., 50.0 49.0 _ _4 ~ 48.0 a 47.0 46.0 45.0 - -. - I I,, 0.0 2.0 4.0 6.0 8.0 10.0 12.0 14.0 16.0 18.0 20.0 Frequency (GHz) Figure 4.3: Characteristic impedance for shielded microstrip on alumina (a = b = 250 mils. W = 25 mils, c7 = 9.79. ) compared to Touchstone. measurements 19] and W = 25 mils, (, = 9.7 ) compared to Touchstone..

Sss modes which sometimes is insufficient. Tlhese features are similar to the behavlor of "qua~.l static" analyses although today's CAD packages generally go beyond "quasi-statics" th employing higher-order, albeit still approximate techniques. The results. of course. are very efficient routines for each component in a microwave circuit which make these packages very powerful and efficient for circuit design. Full-wave techniques have their greatest promise where the other methods fail. for exsample, analysis of high frequencies components, systems with multi-mode interactions, radia tion problems and applications with geometries and accuracy requirements not amenable to simpler analytical techniques, i.e., the type of structures studied here. For instance. Figure 4.4 shows the multi-mode propagation constants for the even modes on a shielded microstrip line produced by a program generalized to handle arbitrary layered structures as discussed. The simple case of microstrip is shown since there is available data for coniparison and verification. The curves overlay the data supplied by Y'amashita [901 with the exception of the dominant mode. Touchstone results are also shlown for the doliiinant mode which are in exact agreement with the present method. It is presumed that the Totichstone results are correct for this mode since the dimensions are not extreme in terms of wavelengths; thus, quasi-TEM assumptions, as used by Touchstone, should be adequate. Most likely, the discrepancy for Yamashita's approach is attributable to the use of pulse basis functions at the strip edge which do not satisfy the edge conditions, although his discretization over the strip improves the capability to approximate the true current. Nevertheless, the form of current assumed with the present method is the exact form for the static case which should be very close to the true solution and therefore is more likely to give better accuracy. Since the exact details of the discretization are not supplied in [901, it is difficult to make a judgement, however, the real point here is the ability to model the higher order modes. (To represent the odd modes, a higher order expansion of the current

89 3.0: 2.0.0 u - Even Mods T — his mahod Daminu. Mode - To,cIuhA. Dan.ma. n Mode - Yamshiw 1.0 0.0 10. 15. 20. 25. Frequency (GHz) Figure 4.4: Multi-mode propagation constants for the even modes in shielded microstrip (a = b = 12.7 mm, h = 1.27 mm, w =.635 mm c, = 8.875).

is needed such as Yamashita's approach or possibly, expansion b) Chebvshev polynomials as in [83].) Besides the obvious utility of this type of information for circuit design and the need for these quantities to be demonstrated in later chapters, the ability to visualize the field or power distribution often provides important insights into why certain structures behave as they do, and also, how the behavior might change when the structure is modified. For example, we later will extensively discuss the special case of a stripline (homogeneously) filled shielded strip) which passes through an aperture in the wall of a cavity. The field distribution in the vicinity of the strip as shown in Figure 4.5 gives a clear indication of the constraints which must be placed on the size of the aperture. One can see that the larger E Field on waveguide cross section Arrow scaled to logarithm of field magnitude 0 dB -d -20 dB-40 dBH Field on waveguide cross section Figure 4.5: Stripline field distribution for the dominant propagating mode.

91 in the wall should be sufficiently large so that minimal energy (proportional to tlhe field is intercepted by the wall. From the numerical data, quantitative criteria can ea,ily l, established. Similar examples for microstrip and suspended microstrip can be found in Figures 4.6 and 4.7, respectively. The upper part of the structure in Figure 4.7 has the same dimenE Field on waveguide cross section H Field on waveguide cross section Arrow scaled to logarithm of field magnitude Arrow scaled to logarithm of field magnitude 0 dB -- O dB - -20 dB - -20 dB - -40 dB - -40 dB - 4 t - in the substrates, one can appreciate how one structure behaves quite differently from the other. Consider for example, what would happen if another strip is introduced on the substrate in these two cases. Clearly, the suspended microstrip field distribution would be more greatly disturbed than the conventional shielded microstrip, and therefore the

92 E Field on waveguide cross section H Field on waveguide cross section Arrow scaled to logariLhm of field magnitude Arrow scaled to logarithm of field magnitude 0dB - OdB -20 dB - -20 dB - -40 dB - 40 dB -. t f i % q f I', i s' t 9 I I I "' I - _ 1 //4 ^- 1X\ J I_ _// f' _ _/ - I \ _..'' _ Figure 4.7: Suspended microstrip field distribution for the dominant propagating mode.

93 one of the benefits of this type of analysis: enhancing our intuitive understanding of the behavior of these structures as well as providing quantitative information. 4.2 Application to Multi-layered Slotline A similar process can be used to evaluate slotlines or coplanar waveguide. Since coplanar waveguide can be treated as coupled slotlines by a simple extension of the method. only the single slot case will be discussed here. We assume the magnetic current is of the same form as in the electric current case which then is valid for narrow slots: 2b(z' - d) (z' y', z) = ( - d) e' - < W/2 (4.30) irWi/1 - 4 (1)2 The variable k9 here represents the set of characteristic complex phase constants associated with shielded slotline modes. They are the eigenvalues of the boundary condition equation: H~t-Ht =3JJJ[GmKGmK]KdV' =0 (4.31) which enforces continuity of the tangential magnetic field ( Hi) in the slot. As before, a particular value for k, is found first, before the fields of a particular mode are calculated. The procedure is the same as before except that the field expressions are changed to reflect the change in the source terms and complementary boundary conditions for the dual fields. 4.2.1 LSE Modes Expanding the VWFs we then find that the LSE mode fields in each layer can be written in the forms: 1 (2 - Lk) ] ELSE =M AS — [i(k2 - + k )] [ -yk: cos k,x cos kzx' + iijkzk. sin k.z cos kzx',i La(k2 + k2) 1 cos,(ci- cosk,(c, - Z) [, + jtan k,(ci - )] (i O) cos k(c - d) cos k,i(z - di) [~eLi - j tan k,i(z - d,)] (i > O) (4.32)

94 HLSE += A, [(2- m,)w(okzj] r~kky sin k-z cos k,z' +'jk2 cos k xcosk { cos kz,(c -z) [4u: tan kz:(cs - z)-j] (i < 0) cosk~~(c~ - d,) ~ (i > 0) cos kz,(ci - di) cos k,i(z - di) [~icL. ta. k,i(z - d) + j] ( ) ( I cosk(z - )z[deL tan k,(z - ) + j] (i > 0) -Ai[ (2 -' )w [ik' cos kx cos k x'] cos 1 k,i;{ cos k,i(ci - z) [i,ui + j tan kzi(ci - z)] (i < O) coskt,(c, 2-dc) tcos k,(z- d,) [il,L. - j tan k,i(z - d,)] (i > 0) (4.33) where i = 0 for the layer with the "slot so that Ao = eijuo + j tan kzo(co- do) (4.34) Al =.5El A1 = T~eL -j tank,l(cl -d (435dl)) By matching the tangential components of the field at the interfaces above and below the strip layer the remaining coupling coefficients are found to be A, = J iAt.lkzik2 cos k,(..1)(cijl -di-d)[,ei t> k,(cj d) + j ( 1) i- 2(i z 0) IAi_,.k_ cos k5(, _)(c,l - d+i_)[f,. tan k,(ci - di) + jJ kz(i+)) - (i+2 cos k,(i+l)(ci+i - di+ )[&.Ui tan L-.i(cj - di) - jl (4.36) The expressions for power density and reaction of the LSE modes are P=ErH; = IA 12(2 - 6m)2WE r k ] k; ] coskx [fcos kz']2 2a2 (k2 + k)f, (kZ)2i Icos k,i(c, - d,)l2 J I'vicos kki(c, - z) + j sin k (cj - z)12 (i < 0) t Li cos kski(z - d) - jsin k,i(z - di)l2 (i > 0) (4.37) a2 (kx + k2),] Lk? co02 k,,(Ci - di) { [ui cos k,i(ci - z) + ji sin ks(c, - z)]2 (i < O) [cti cos k,i(z - di) - j sin k(z - d)l] (i > 0) (4.38)

4.2.2 LSM Modes Similarly, the LSMI mode fields can be wvritten as ELSM B, []Yk2 COS k,x COS k - yjk1k, sin krI cos kx 1 { os f kZ,(c, - Zk) ['mUZ + j tan k, (c, - z)m (:, 0) cosk,(c, - d,) cos k,(z - di) [mL, - jtan k,(z - d,)] (i > 0) ~B, [(2 - b] [EkB sin -kx cos,kz']. acik,i 1 O cos k(z(c, - z)[muitan ki(c,- z)- ) - j] < 0) cos,(ci -i) Los) ok,,(z -d L i,) z[.Wt,Ld tan k,,(z - d,) + j (i > 0) 1 f cos kl,j(c, - z) [imU, tan kz,(c, - z) - jJ (i < 0) (4.40) where B = -O (4.41) 7muo + jtaii ko (co - d o) cosk(| l - ) { ( cos..i(ci- ) [i-;litank,i(ci - z)- + ] + os ki:l)i(Z - di) [/, t, i tan k~i(c - di)+ (i > O ) (4.42) Bm = -i tankl. k(co - di) I B, I jkz, B,= k:(, —) cos k..(j_l)(ci-i - d,-,)[(/~mLi tan kt,(c, - d,) + j} i > 1) kz(l+1) cosk,(ig~l)(ci+l - di+t)[,mu, tank.,(c. - id) -i (i < 0) (4.43) For the LSM modes the required terms for calculating power and reaction are 2(k2 + k2) 1-4k1 2k,J Icosk,-d,)t2 { IT,mUisin k:,(Ci- z)i j c osk,(ci- z)k2 (i - 0) (4.44)

96 B(2 - m)2wEo kr kky sin krl (Lfcos k,z'1 R = -E, ( k0H) x(-kv ) B, a2 [ tkz][ kz-(k2 + k2)] CoS2 kzt(c - di) { [|m,u sin kz;(c -z) - j cos k,,(c, - z)}2 (2 < O) [mL, sin kz(z - d) + j cos k,;(z - d)]2 (z > 0) (4.45) 4.2.3 Cross Terms There are also two sets of cross-terms from the ELSM x HLSE product. The power terms are PN- A -_e kr 1 kl i cosZ kIx [f cosk,z'I2 = r 2- i 2a (k + k2) (k )2 I os k,(ci - d()12 f [,,mucos kz(c, - z) +' sin kz,(c - z)] ([cu, cos k,i(c - z) + jsink,,(c, - z)]' (i < 0)' [mL, cos k,,(z-d d)-j sin k,,(z-di)]l [,cLi cosk,i(z-d d) -j sin k,,(z-d,)' (i > 0) (4.46).(2-6m)2Wfo r ks ][ kk; sin2 k:z 4 cosk.,x P = _ -E. = B,A — a2 k.,ic k;,(k2 + (k;)2)] cosk, -(c d-)l2 ([4mu, sin k,(c - z) - j cos k,,(c -z)] ([uj sin k,-(c, - z) - j cos k,( z)]c (- < O) 1 [T,mL, sin k,i(z - d) + j cos k,i(z - di)][,Li sin k,,(z - d ) + j cos k,,(z - d)]' (i > 0) (4.47) Similarly, for the reaction we have (2 - bm)2W[, k2 1 v 1 Cos0 2k: L cos k.z'l2 R = E:(kv)fHz(-AkV) = B.Ai(2-6m) [ i] [J cos2 k_ (c -O ) { [(mu cos kzi(ci - z) + j sin k,i(ci - z)] [ cos k,i(ci - z) + j sin k,-(c - z)l (i < 0) [im Li COS k,i(z - di)-j sin k,i(z- di)l [iL, cos k,i(z - d ) - j sin k,;(z - di ) (i > 0) (4.48) R = Eg(kv)Hx(-k) = BjA, (2 T6n)2WC, [ k 1 [ k:ky 1 sin2 k_x [f cos k,z')2 vR~~t va2 ki,cI i[ k8i(k2 + k )] cos2 k,i(c, - d1) { [mUi sil k:,(ci - z) - j cos k.(ci - z)l [ui sin k,i(ci - ) - j cos k,(c, - z)] (i O) [7m.L; sin k,i(z - d) + j cos k,,(z - di)] [fL; sin k,i(z - d) + j cos k;,i( - di)] (i > 0) (4.49) The integration of these terms on the cross section involves the same forms as for the

97 electric currents which have already been given. 4.2.4 Application to Finline and Suspended Finline As with strips. the distribution of the field provides insight into the behavior of tlvl structures. Two examples illustrate by showing the change in the field structure for finiine without a substrate compared to tlhe case where a dielectric substrate ((r = 2.2) is added. These cases are shown il Figures 4.8 and 4.9 E Field on waveguide cross section H Field on waveguide cross section Arrow scaled to logarithm of ficld magnitude Arrow scaled to logarithm of field magnitude 0 dB - 0 dB - 20 d - -20 dB - 40 dB - -40 dB - iA i ~,.W' o o X s t' 4' "44 I I I Ii i 4 I/ / / \ \ Figure 4.8: Finline field distribution for the dominant mode.

98 E Field on waveguide cross section H Field on waveguide cross section Arrow scaled to logarithm of field magnitude Arrow scaled to logarithm of field magnitude 0 dB -- 0 dB -- -20 dB - -20 dB -40 dB - -40 dB - I I I_ I I, i _ _ __ __ _____I 4 I 4 4 I t I I ~ 4 - 4 111141 4 _ I d _ i / / Fge.:uee/nfedsruofrhdI \\\\ // I * 4 I I I e I e I Figure 4.9: Suspended finline field distribution for the dominant mode.

4.3 Scattering from Pins in Rectangular Waveguide The insertion of wires or pins thlrouglh tile ground planes is a teclhnique used for manv years to suppress parallel plate waveguide modes in stripline and for structural support. It is also a convenient way to form waveguides, cavities or isolation walls in stripline circuits. The experimental work discussed in Chapter VI is an example of thle formation of a cavitv where our particular need for this analysis will become evident. Other applications include the use of posts (or strips) in a rectangular waveguide as reactive elements in filter and matching networks [9, 10, 6S] and for active device mounts and tuning elements [(7 24j. These types of applications have been the focus of most of the tlieoretical and experimental work appearing in the literature. In contrast, what we are interested in here is the effectiveness of a'wall of wires' as a short circuit. Analytical techniques in early works are primarily based on variational methods [54. 48, 45]. These approaclhes, however, become impractical for more than a few posts or posts with irregular spacings and are limited in accuracy at high frequencies. Even later works focussed primarily the study of thlree posts at most with emphlasis on accuracy for post of relatively large diameter [43, 44, 3]. The emphasis on a limited number of larger posts is the opposite of what we require here where we wi-ll tend to use wires of small diameter and of greater number (and density). It is also evident that larger posts require more complex formulations and numerical treatments than is desirable for our situation'. Image theory or grating formulations as in [29] or especially [46] would seem to be appropriate for this problem but appear to be excessively complex and have limited potential for combination with other structures. The approach developed here will therefore be in spirit of the approach given in [43] which offers a good balance of simplicity and flexibility. In addition,'Unles one already hu a numerical code to treat the problem at hand, it is always desirable to use the simplest formulations which can produce results of sufficent accuracy and therefore can be quickly implemented. Another consideration is whether the resulting numerical model can be efficiently evaluated which usually, but not always, favors the simpler formulations.

100 the method will be even furtlier simplified for wires of small diameter 4.3.1 Reflection Coefficient Formula Derived from the Reciprocity Theorem Consider a grid of wires appearing in the cross section of a rectangular waveguide as illustrated in Figure 4.10. Let us assume that the diameters of the wires are sufficiently z / at the center of the wire, i.e., i 0 Figure 4.10: A grid of vertical wires in rectangular waveguide. at the center of the wire, i.e., J = i,6(z -,)6(y-y,) (4.50) Furthermore, the waveguide dimensions are deemed to be suclh that all modes but thle dominant one are cut-off. The current on the wires is also assumed to be excited by the dominant mode field which has no Z variation. This fact justifies our assumption that the current also has no z variation. For multilayered structures this assumption would no longer be valid and the formulation must be made more general, however, it is sufficient here since the present need for this effort is restricted to stripline in support of Chapter VI.

101 We can derive a reflection coefficient representing the scattering of the dominant ml(im by an application of the Reciprocity Theorem [33] in the form E,,x -Ho 6- 6x = Eb 6J- E - J~ dl The locations of the surfaces defining the volume of integration are coinciderit with thip side walls of the waveguide and two transverse planes on either side of tile grid of wires The transverse planes are assumed to be far enough away from the grid wires so that only the dominant mode has significant field strength. Now let the Ea and Ha fields represent the normalized fields of the dominant mode, incident from along the -y axis, hiving no source terms within the volume of integration, i.e., Ja = 0. Then thiese fields are given by Ea = El = sin(kmz)ekYY (4.52) Ha -HI = -i: k sin(k,z)e-{ 3 where km = m7r/a, m = 1 for the dominant mode and ky = x/k2 - k. The Eb and Hb fields will be produced by the currents on the wires and radiate in both tle +y and -y directions. If the currents on the wires, I,, are those which are excited by the above incident field, then we can write Eb = rElo = i rsin(kmz)e+J)k' (4..54) Hb = rHio = -i r k sin(kmz)e+JkY (4.55) where the top sign is for y < y' and the bottom sign for y > y'. Equation (4.51) nowv reduces to f if E,o J dV r = - % _ TJ. Id (4. 56) 2ff Eo x Hl1 y *dzdz Note that to obtain this result, the surface integrations over the walls of the waveguide were found to be zero, since the tangential E fields are zero there for perfectly conductlne

102 walls and also the surface integral over the waveguide cross section on the +y side of tihe wires evaluates to zero. The remaining integrations are readily performed to give r = _ 2rfu 2 Z Ij sin(rx,/a) (4.57) aVk/.2 - (r/a)2 in We thus need only determine the unknown currents from which the reflection coefficient is produced by this simple summation. 4.3.2 Method of Moments Formulation In the previous section, the assumption of constant current on the z dimension hlas already been stated. In the parlance of the Method of Moments this is a'pulse' basis function which in effect has reduced the problem from three dimensions to two. For small diameter wires as treated here, the problem can be further reduced to one dimension, if the grid of wires are all contained in a transverse plane, however, this produces no significant ad vantage. Since we have previously found the dyadic Green's functions for homogeneously filled rectangular waveguide (section 2.4), we cali use Equation (1.33) to derive the electric fields in the waveguide. With the current as stated by Equation (4.50), the z component of the E field produced by the j't current filament is given by: E, = -w la y k sin(k,z) sin(kmz )e-')kV*IY-YI (4.58) a M=1 Ik It remains then, to find the set of Ij's which produce a total electric field that satisfies the boundary conditions on the wires, specifically E, + i * Elo = O. The simplest way to enforce this condition is to use point matching on the surface of the wires which is to use a delta weight function. Although found to produce identical results, a two point matching scheme was used to provide symmetry balance in which the field is taken as the average

103 over two points of the wire surface. Tile resulting expression for the field is E -= - I, E A sinlk(z i r,)j sin[km(, ~ r,)l sin(kz, ) a k wuhere i is an index indicating the field evaluation points and r, is the radius of the zt' -'r, The ir, notation is taken to imply the averaging of the field contributions at these point" The combined field expression can be written in matrix form as [Z,l [Ij] = [ - Elo, (1 which is solved by matrix inversion for the unknown currents. 4.3.3 Validation In order to investigate the validity of this formulation we first compare to the data of Marcuvitz [48] for single posts. The accuracy of this data is weUL established for diameter to waveguide widthl ratios (d/a) of up to 0.25. The equivalent circuit for thle siingle post is as shown in Figure 4.11. From this equivalent circuit we can convert the values given Xb Xb Port Xa Port 1 2 Figure 4.11: Equivalent circuit for single post in rectangular waveguide. by the formula in [48] to a reflection coefficient which can be compared to the results of Equation (4.57). Since our formulation is a simplified version of the method fou nd in [43].

104 1.0-.. ___ - -, - -- 170.,-,., - -,- - "'" 9 ~b160. 0.8 150. C...o 0.7.- 140. U 0.6 - - C 0,c 05,c, 120. 0.4 510...t 0.00 0.01 0.02 0.03 0.o 0.05 0.06 0.07 0.08 0.09 0.10 0.00 0.01 0.02 0.03 0.04 0.05 0.06 0.07 0.08 009 0 10 Normalized Post Diameter (d/a) Normalized Post Diamncter (d/a) Figure4.12: Magnitude and phase of reflection coefficient for a single conducting post in rectangular waveguide. Point match Method of Moments (MoM) results compared to thle data of Marcuvitz [48] (A/a = 1.4). we would not expect the range of its accuracy to extend as far; thus, we will limit the range of our discussion to dla < 0.1. Moreover, we are specifically interested in thin wires with diameters (d) much smaller 0.la, on the order of 0.02a, the approximate value to be used later in Chapter VI. The results over this range is given in Figure 4.12 which shows the magnitude and phase of the reflection coefficient of a single post in rectangular waveguide as compared to Marcuvitz's data. As can be seen, the comparison is excellent with a maximum phase error of less than four degrees for the largest wire diameter. Thus, we have a first indication of the accuracy of the method. Before proceeding to further results, it is instructive to further examine the single post case. The behavior of the element values of the equivalent circuit foUow the curves illustrated in Figure 4.13. By evaluating the input impedance for one of the ports with the second port matched and plotting it on the Smith Chart as in Figure 4.14, we see that the input impedance progresses from inductive to capacitive as the diameter is increased. We also note that the single post makes a surprisingly good short circuit at a_diameter as

105 0.80.'...... 0.70 1 0.60 _ \ Shunt inducLve Rewaanc X. XJa = 1 4 \. —----- Scncs Cipaciuve Re.aanc X ja 4= 4 0.50 0.40 0.30 c- 0.20 0.10 0.00 0.05 0.10 0.15 0.20 0.25 Normalized Diameter ( d/a ) Figure 4.13: Behavior of element values for the equivalent circuit given by Marcuvitz [481 (A/a = 1.4). small as d/a = 0.1.5, the diameter at the 180~ phase shift point. Not surprisingly, as more pins are added the wires form an even better short circuit producing ever higher shunt susceptance values. For example, Figure 4.15 shows how the shunt susceptance value increases rapidly as the number of equally spaced wires is increased. In this case we have lumped the reactive behavior of the grid into a single shunt element on the transmission line, i.e., replace the series capacitors of the equivalent circuit of Figure 4.11 with shorts and the shunt inductor with a general reactive element. (Note also that all wires in this investigation will be located in the same transverse plane of the waveguide whiclh is the reference plane for thle equivalent circuits.) It is interesting to observe that the normalized pin diameter (d/a), where the susceptance crosses over from inductive to capacitive, depends on the number of wires and generally moves toward smaller post diameters as the number of posts increases. This suggests that for a given

106 Figure 4.14: Input impedance as a function of post diameter with a matched load port. Post diameter increases in the counter-clockwise direction from d/a =.005 to d/a =.25 in.005/,steps, demonstrating the transition of the input impedance from inductive to capacitive.

107 200.0: — i —-, - T --'X - - 200.0 i, OneC Pa \ 150.0 Thr Poss L s 100.0 ------ FtPO, - -_ —-' — Sve. Pc as 50.0 -. ----— 0.-_ -. -50.0 - -.0,'. V, -150.0 -200.0 0.000 0.020 0.040 0.060 0.080 0.100 Normalized Pin Diameter ( d/a ) Figure 4.15: Shunt susceptance behavior for up to seven wires placed evenly spaced across waveguide cross section. number of wires equallv spaced across the guide, there is an optimum post diameter whicII most nearly produces an effective short circuit. Similarly, it implies that additional posts do not necessarily improve the capability to simulate a true short circuit at the reference plane. With these large shunt susceptance values, the real part of the input impedance becomes negligible so that the network can be treated as a one port terminated with this reactive element. As such, the one port can also be modelled as a true short circuit located at a distance from the reference plane determined by the calculated value of susceptance and the transmission line equation: B = cot(Pl) (4.61) where B is the susceptance, I is the distance from a true short to thle reference plane and: = 2r/A9, with )g the guided mode wavelength. The argument /1 is the phase angle of the reflection coefficient for the line. The accuracy with which we can determine this angle

108 is a measure of lhow well the grid of wires is treated by tlhe simplified model. tilus bc-,r;::. the focus of the experimental results and comparisons to follow. To further confirm the accuracy of this method, measurements were made with ai; HP8510 Network Analyzer on various combinations of wires in X-band waveguide (0.4 x 0.9 inches) at 12 GHz. A series of seven equally spaced holes where drilled in a line on the broad wall, transverse to the axis of a section of waveguide. The holes themselves were small enough in diameter and sufficiently spaced so that they did not significantly perturb the propagation of the fundamental mode, a fact verified by subsequent measurements. The reference plane was established and the fixturing de-embedded by performing a oneport calibration with two short circuited waveguide sections of different lengths and a precision waveguide load. To make the measurements, different patterns of wires were inserted through selected holes and the reflection coefficients recorded. The patterns used will be denoted by a series of ones and zeroes, for example the case of the single centered post would be designated by the pattern'0001000'. The results of one set of measurements are shown in Figure 4.16. In this and the following charts, data marks correspond to the three wire diameters used (d =.025,.033..039 in.), the filled-in marks representing the corresponding measured points. As can be seen by drawing a line from the center of the chart through the various points, the accuracy of the predicted phase angle is quite good. Similar results are found for wires in pairs as shown in Figure 4.17. Figure 4.18 is an experimental demonstration of the case discussed above where tle number of pins is progressively increased. This is perhaps the worst case where errors in phase angle are on the order of five degrees for some points. Some of these errors are attributable to experimental error due to variables such as mis-alignment in the transverse plane and tilting of the wires in the holes, which necessarily must be larger to accommodate

109 ~\~'~ x x x) —--- P: ln. OOiOv/ Figure 4.16: Reflection coefficient measurements for a single post at various offsets from the centerline compared to predictions from the simplified model. Wire diameters (d =.025,.033.039 in.) increase in the counter-clockwise direction.

110 \ P-laom - I!~ Figure4.17: Reflection coefficient measurements for pairs of wires at various offsets Fiure4.17: Reflection coefficient measurements for pairs of w ires at various offsets from the centerine compased to predicted values. Wire diameters (don..025,.033,.039 in.) increase in the counter-clockwise direction.

111 ~\'~~~ X V X — o —.~ Prom l001 00 -- m, —- P, ~ttan 01 I 1 I 10 Figure 4.18: Measured reflection coefficients for centered wire grids of equal spacing and increasing number. Wire diameters (d =.025,.033,.039 in.) increase in the counter-clockwise direction.

112 different diameters. Nevertheless, the overall results suggest that the method is capable of predicting the phase to within a few degrees. The final plot, Figure 4.19, is an illustration of a case very similar to one which will -,l v —- Pum.m:1101111_ PatemIlOI 1101111 Figure 4.19: Measured and predicted reflection coefficients for wire grids with a single pin missing in the sequence. Wire diameters (d =.025,.033,.039 in.) increase in the counter-clockwise direction. be encountered later. In this instance, an evenly spaced row of wires were installed across the waveguide cross-section, however, one pin was removed at various locations in the sequence. As will be seen in Chapter VI, this type of approach is used to allow a strip to pass through the pin curtain. The ability to predict the reflection coefficient for this case

113 is very important for certain situations as will be seen. Although the comparison for i. case looks reasonable, it is pushing thle limits of the experimental errors inolved h it i fixture. However, a consideration of the previous results leads to thle conclusion tihat'tle results are reliable. 4.4 Summary and Conclusions In summary, we have seen how problems with uniformity in one dimension can tbe arallyzed with a high degree of accuracy. The uniformity allows the structure to be represented by a single modal series which can be numerically evaluated efficiently. The use of impedance boundary conditions and a LSE and LSM expanision of tlihe fields greatly simplifies the modelling of multilayer substrates and superstrates. Thils particlilarl\ facilitates the computation of fields throughout the structure since most of thle evaluiation can be done analytically. Application to shielded strip and slot geometries icelds 1,ot 1 reliable quantitative information as well as visual representations of the fields providldma insight into the behavior of the structure. A simple model for scattering from wires in rectangular waveguide has also hbeen developed. The simplicity of the model is in contrast to methods discussed in the literatiire which are oriented towards fewer posts with relatively large diameters. The present method. however, emphasizes posts of greater number and smaller diameter. Experimental results were obtained to verify the modelling ability of the technique. The need for this model in the context of the present work appears in Chapter VI.

CHAPTER V COUPLING THROUGH STRIP-FED SLOTS A class of structures which couple from one guiding structure to another through an aperture is important in microwave circuits, aside from thle radiating slot. In fact, for antennas, the coupling of waveguide to waveguide through a narrow slot in a common wall has been used for many years in the design of corporate feed networks for slotted waveguide arrays. As discussed in Cliapter 1, the case of coupling between shielded strips in a similar manner is becoming equally important in current and future systems, including similar feed networks. In this chapter, the analysis of these types of couplers will be presented along with experimental verification of the numerical results to demonstrate the applicability of these techniques to microwave and millimeter wave devices. 5.1 Network Analysis The basic structure of the coupler to be discussed is as shown in Figure 5.1. Variations on this geometry include cases with microstrip lines on the same side of the slot; multilayered substrates/superstrates; reverse couplers where the lines exit on the same wall; additional parallel slots and lines; and 3- and 4-port networks, among others, but all can be analyzed using the same approach. Integral equations have been formulated for this problem in Cl4apter I and are solved as outlined in Chapter III for currents on the slot and strips. Therefore, in-this chapter 114

115 Figure 5.1: Geometry of basic coupler. we need only discuss the interpretation of these currents in order to extract the network parameters which characterize the coupler's behavior. Solutions for the currents are found with even and odd gap generator excitations at the line ends. These are sufficient to characterize the two-port problem since all excitations can be decomposed into even and odd components. The approach can be generalized using V-port network analysis [14, pp. 157-158] but this aspect is beyond the focus of this work. From the even and odd currents on the microstrip lines, even and odd impedances are found by measuring the relative distance (d) from a standing wave maximum to the location of the slot as illustrated in Figure 5.2. We can also estimate the guided wavelength for the even and odd modes from the standing wave patterns by measuring the spacings between minima or maxima. If the materials are lossless, as we typically can assume for most microwave circuit applications', the expression for the reflection coefficient referenced'For slightly lousy structures, the estimation of the attenuation constant can be easily determined from the ratios of successive current maxima or minimai55]. For larger louses the situation is more complex, since not only the envelope of the standing wave current but also the spacing between maxima and minima vary along the line [27, sec. 5.4]. In this case one must resort to a parameter estimation scheme to find the

116 current magnitude opcn end -. d _' slot posiLion Figure 5.2: Measurement of location of current peak relative to the slot for standing wave calculation. to the slot reduces to F = -eJ4Wd/A (.5 1) which produces an impedance according to. z i+r (5.2)!-r The even and odd impedances are then combined to form the Z-parameters which, in the symmetric case studied here, are given by the simple expressions: Z-l = + = Z22 (5.3) Zo - Z, Z2= Z Z12 (5.4) Finally, the even and odd impedances may be combined to produce S-parameters tlrough the transformations: ZS 1 21 Z+ 2Z1 = S2 (5.5) S21 = Z: + 2z- Z l + 1 = S12 (5.6) optimum fit to the current with the attenuation and propagation constants as parameters.

117 which are used to characterize the coupling behavior. Before proceeding, the limitations of this technique should be noted. Transmission line analysis of the current in this way assumes that there is only one propagating mode on the line at some distance from the discontinuities. For the shielded structures treated here. this requires that the cross-sectional dimensions of the feeding microstrip lines be such that all higher-order modes are cut-off or, looked at another way, a given cross-section restricts the maximum frequency for which circuit parameters can be reliably produced with this technique. Also, in order to estimate the phase constant, the line must have a minimum length, typically on the order of the wavelength. Moreover, for good numerical stability of the results over a wide range of parameters, it has been found useful to average a number of estimates on a line approximately thlree wavelength long, discarding the maximum and minimum values. This restriction demands more unknowns and hence greater computation time, however, the results are quite reliable. The alternative method based on tile Reciprocity Theorem developed in Chapter VI, overcomes these limitations at the expense of a more complex formulation. 5.2 Fixture Design To verify the results, we have designed and constructed the fixture shown in Figure 5.3. Sample substrates with various line and slot dimensions are installed in the fixture in different combinations to allow frequency response measurements. A number of circuit boards were made: One set of boards was double-sided with a microstrip line etched to a certain length relative to a slot etched in the ground plane on the opposite side. The second set was one-sided boards designed to be held against the boards of the first set by the fixture, with microstrip lines of corresponding lengths. The Lines are excited by Eisenhart connectors which provide a reasonable match over a broad frequency range, even when the contrast in (, is high (e, = 1 for the connector, ~, = 10.6 for the substrates). In our case

118 Figure 5.3: Photograph of coupler fixture assembly. this is partly a result of the good match in dimensions between the boards and thec tip to outer conductor spacing of the connector. Another critical factor is the transition fronl the outer conductor of the connectors to the ground planes of the boards which must be carefully designed. The match at the connectors was optimized to minimize the launching discontinuity, allowing us to avoid the necessity for TRL [341 or similar de-embedding schemes which require additional and more complex fixturing. 5.3 Numerical and Experimental Results A coupler with the geometry of Figure 5.1 was analyzed using the above techniques. The parameters which can be varied in this design are numerous, consequently, only a few variations will be presented here. In all cases, although not required in general, symmetric geometry was maintained to simplify the even and odd mode analysis, as discussed above. Also, in all cases the cross-section for the cavity was 0.25 x 0.25 inches; the substrates were 0.025 inches thick with E, = 10.6; and the slot and line widths were 0.025 inches. The

119 cavity length was fixed at 2.0 inches for thle measurements and varied for the numerical results to allow an approximately constant 3A feed-line length. This dimension does not affect the results since for all frequencies considered here, the cavity is below the cutoff frequency of the higher order microstrip modes and the reference plane was fixed at the location of the slot. To illustrate the behavior of the coupler, we first examine the influence of various parameters at fixed operating frequencies. The effect of the line stub length (I) is shown in Figure 5.4. It can be seen that the stub is initially too long for an ideal match at this -3.0 7 9 -9.0 10 71.,0 i-8.0 -21.0 0.000 0.025 0.050 0.075 0 100 0.125 Line Stub Length (inches) Figure 5.4: Effect of the line stub length (1) on S21 and S11 magnitudes (s = 0, L, = 0.25 inches and f = 12.0 G Hz). frequency. However, as the stub is progressively shortened, a certain length "matches" the two port coupler and with further shortening the match gets progressively worse. We can interpret this effect by examining the equivalent circuit shown in Figure 5.5. Variation of the stub length has the effect of changing the position of the current maxima (virtual shorts) and minima (virtual opens) on the lines relative to the slot, thus varying the degree of coupling through the slot represented by the coupling transformers. Cons-qeqtently, the

120 open /, Port end // jX2 Port,X open 1 e j |cend! jx. Figure 5.5: Equivalent circuit for a 2-port coupler. peak coupling occurs when line stub length places a current maximumn below the slot or lengths in odd multiples of,, A/4. The opposite effect occurs when the line stub is approximately in multiples of A/2 in length so that there is a virtual open circuit beneath the slot, in which case there would be very little coupling between the line and slot. A similar effect is observed for variations in slot length (L,) as illustrated in Figure 5.G. Again using the transmission line analogy, one can interpret this effect by transforming the impedances at the ends of the slot to the center. These end impedances are nearly short circuits, the difference being due to fringing fields which extend beyond the ends of the slot line, fully accounted for by the full-wave analysis. At the matching length, the resulting transformed reactances at the center cancel the reactance associated with the junction, thereby matching the two ports. As the slot becomes very short, the field in the slot is effectively "short circuited"; thus, coupling is reduced. S21 then tends to zero while SI1 approaches unity (since the structure is closed and assumed lossless). All of these effects would be expected to repeat as the slot length increases in multiples of A, however, for the case studied here, the maximum slot length is limited by the dimensions of the shicld,ng

121 -.3.0' -9.0 _ * 2 -7.0. -15.0 -18.0 -,o',' -21.0 0.000 0.025 0.050 0.075 0.100 0.125 0.150 0.17S 0.200 0.225 0.250 Slot Length (inches) Figure 5.6: Effect of the slot lengthl on S21 and SI1 magnitudes (s = 0,1 = 0.049 inches and f = 18.0 GHlz). package which have been chosen to allow propagation of only the dominant microstrip mode. To generate a frequency response, the programs are run at each frequency of interest and the slot and line lengths are varied to form a parametric database. The database is tlhen scanned and interpolated to assemble frequency response plots as functions of the geometric parameters. Measurement of one of the assemblies is shown in Figure 5.7 in comparison to corresponding numerical results for a stub length of I = 0.115 inches. The position of the high frequency corner of the response was found to be very sensitive to the length of the line stub. As discussed above, this corner is controlled by the length at which the stub is approximately A/2. Since the effective dielectric constant for the microstrip is approximately c,.ef = 7.8 at 17.0 GHz, a null is predicted in the coupler response for that neighborhood, in good agreement with the results shown. The error bar on the high end indicates the sensitivity of the high frequency corner to a f5 mil error in line stub length which is well within the expected tolerance-errors for positioning the stubs rclI;tt

122 0.0 I. -3.0.6.0 -9.0.12.0.o i..,. -..21.0 0.0 3.0 6.0 9.0 12.0 15.0 18.0 Frequency (GHz) Figure 5.7: Comparison of theory and experiment for S21 magnitudes with s = 0,1 = 0.115 in., L, = 0.250 inches. The error bar indicates the influence on the high frequency corner of a ~.005 inch change in stub length. to the slot. In view of the above, we conclude that the theoretical results are in excellent agreement with the experimental data. In fact, we were able to move the upper board slightly toward the slot to extend the stub length somewhat, shifting the high corner to a lower frequency as expected. However, this also created problems with the match at the Eisenhart microstrip launchers so these results are not shown. The'sidelobe' which can be seen at the high frequency end, is also attributed to tolerance errors for the line stub lengths. A difference in lengths would produce multiple nulls in the response at the high end resulting in undesirable sidelobes in between. Because of the high sensitivity to line length, owing in part to the high dielectric constant, the amplitude and span of the sidelobe is a strong function of the relative line stub lengths, a fact which can be observed when the boards are slightly shifted as described above. The sidelobes do not appear in the theoretical result since a difference in stub lengths between the upper and lower lines introduces an asymmetry which has not been included in the

123 current numerical model and contradicts some of the assumptions stated for the network analysis. A more general model can be produced by extending the network analysis to tihe non-symmetric case. The ripple in all the measurements can be shown to result from mismatch at the microstrip launchers. The measurements are particularly sensitive to this connection because of the high dielectric constant of the substrate. Ripple occurs to various degrees throughout the measurements and is also influenced by small air gaps between the connector assembly, fixture and substrates. It could be removed by more sophiisticated de-embedding techniques, however, this requires additional fixtures. Nevertheless, the ripple shown in the results presented here is not substantial and does not significantly interfere with the fundamental behavior of the devices. Also, the broadening of the low frequency response is typical in the measurements. We were not able to identify a direct cause for this effect, however, we suspect that it is related to the fixture/connector interface, since we have not de-embedded these transitions. We also postulated that some of the anlomalies might be caused by the side-wall grooves in the fixture which hold the double-sided board in place. This possibility was eliminated, however, by installing movable side-wall shorts which are visible in Figure 5.3. The remaining discrepancy is perhaps a slight additional loss found in some of the measured results. To de-embed the losses for the structure, a through line was measured and the remaining measurements were post-processed to compensate for conductor and dielectric losses on the microstrip lines. This process does not correct for losses associated with the slot including both conductor and dielectric losses, additional losses on the cavity walls, and losses due to the added line lengths. The remaining differences are attributed to these factors together with measurement errors and are judged to be within acceptable limits.

124 Measuremnents on a different line stub length are shown in Figure 5.8 again slouw.inr 0.0 -3.0 -6.0 -9.0 -12.0 -15.0 M- wd-- -- lbwl y 18.0 -21.0 0.0 3.0 6.0 9.0 12.0 15.0 18.0 Frequency (GHz) Figure 5.8: Comparison of experiment and theory for S21 magnitudes with s = 0,I = 0.080 in., L, = 0.250 inches, illustrating the control of the high frequency corner with the line stub length by comparison to Figure 5.7. good agreement witli the theoretical results. In this case, the shortening of the stub length has moved the high frequency corner out of the range of the measurements. What is interesting to note about this case is that the low frequency corner of the response is qllite insensitive to this change in stub length. Also of particular interest is the wide bandwidth of this transition. Figure 5.9 illustrates the effect of shortening the slot length. One consequence is reduced coupling in the passband which was also demonstrated in Figure 5.6. We also see in this result, some movement of the high frequency null due to a shortening of the line stub length. The final plots, Figures 5.10 and 5.11, show the influence of the line separation pa rameter (s) on the frequency response. These figures are to be compared to Figures 5., and 5.8, respectively, and in general show a narrowing of thre frequency passbandl ru

125 0.0... I.... ~~~~~~0.0~~~ -3.0 -6.0 -9.0 0 -120. —- lbwy.18.0 -21.0 - - J I.., t.,... -21.0 0.0 3.0 6.0 9.0 12.0 15.0 18 0 Frequency (GHz) Figure 5.9: Experiment and theory for S21 magnitudes with s = 0, I = 0.110 in.. L, = 0. 1.53 inches, showing the effect of shortening the slot length. 0.0 -3.0 -9.60 -12.0 0.0 3.0 6.0 9.0 12.0 15.0 1I.0 F w=uey (GHz) Figure 5.10: Comparison showing the effect of line separation with l = 0.115 in.. L, = 0.2.50 in. and s = 0.125 inches.

126 0.0 -3.0 -6.0 -9.0 -12.0 / I:+ — Tboary -18.0 -21.0 0.0 3.0 6.0 9.0 12.0 15.0 18.0 Frequency (GHz) Figure 5.11: The effect of line separation with shorter line stub length; I = 0.080 in., L, = 0.2.50 in. and s = 0.125 inches. is increased. This is to be expected, since we have now introduced an additional length parameter whlich can influence the response thlrough its relationship to wavelength. Here again, the numerical model is judged to have correctly predicted the coupler behavior after the experimental artifacts are considered. 5.4 Summary The frequency response plots shown demonstrate the utility of the structure as an interconnect. With proper selection of the geometric parameters such as line and slot widths, lengths, line separation, substrate heights and materials, the frequency response can be tailored to give the required center frequency, bandwidth, shape, etc. As has been shown, very wide bandwidths can be achieved which makes the structure very versatile. Avoidance of via-hole transitions and their inherent limitations by the use of planar structures to form vertical interconnects, together with the ability of the model to accurately predict the coupler behavior as demonstrated by experimental results, are especiallyj important

127 considerations for design of monolithlic circuits. In addition, altliougli the approacht dil.cussed here uses certain simplifying assumptions about tie symmetry of the structurars. the technique can be readily adapted to the general case.

CHAPTER VI ANALYSIS OF STRIP-FED RADIATING SLOTS Arrays of slots have been widely used in antenna design for nearly fifty years. Tlc mnost common approach has been to cut slots in the walls of rectangular waveguide formiiig a Linear array. An arrangement of linear arrays, fed by additional slotted waveguides, has frequently been used for two-dimensional arrays. The design and analysis of these approaches has proceeded over many years to the point where accurate numerical techniques are now available so that costly, time-consuming empirical techiniques caln be bypassed 123. 61]. More recently, complex systems have imposed new antenna requirements with denialids for compact conformability, reduced weight, and higher order antenna functions stlch as electronic beam steering, polarization control and power generation. A suitable approachi employing stripline-fed series slots isolated from higher order modes by shorting pins, has been analyzed by Shavit and Elliott [67]. Their approach, together with an alternative coupler feed arrangement, is illustrated in Figure 6.1. The isolation of the slots by the cavity walls, in addition to simplifying the design, has been shown to provide significant advantages for phase steerable antennas [47]. Other variations on this approach may he adapted to fit system requirements including the incorporation of active elemenits and vertical integration of more complex circuits on multi-layered substrates. As indicated in earlier chapters, we extend the analysis to include strips on multilayered substrates and superstrates, as well as to include the effect of a dielectric cover on 128

129 e~ b -~ ~ ~~~~~~~~~~~~ -e ~~~_::a~ a -- ~ ~ ~ ~ ~ ~ ~ ~ ~ ~ _ *.0 - [, I s w.e o II * S~~~~~~~~~~~ Figure 6. 1: Strip-fed cavity-backed slots with coupler-fed and series-fed corporate feed arrangements. U~~~~~~~~~~~~ 0 Ii~~~~~~~~~~~ S::__ Figure 6~~~~~.: Sti:e:aiybce lt ihcule-e n eisfdcroaefe r rangements.~ ~

130 the slots. As will be seen the present formulation substantially improves accuracy. to tlie point where one could expect to perform the design without empirical methods. 6.1 Equivalent Slot Impedance In order to produce a slot whose resonant length is less than the width of the cavity, the cavity dimensions must be such that the first higher order mode in the cavity cross-section - the TE1o mode - is not cut-off. In this case, since more than one mode can propagate on the microstrip line, the method used to extract circuit parameters in Chapter V cannot be used. In addition, these cavities are typically on the order of one-half wavelength in length so that the lines would probably not be long enoughl to get reliable circuit parameters using the Standing Wave Method. The scattering parameters for the slot can, however, be derived through application of the Reciprocity Thleorem [331 to the cases illustrated in Figure 6.2, resulting in slot position H H TEM a) E1-M H J waveguide aperture slot * position H K waveguide apetnure Figure 6.2: Two cases for application of the Reciprocity Theorem.

131 JlJagd [E~TEM X HA - E X HTEkf dS = JJO HTEf h; dS G6. waueguade lot where we have assumed the incident wave is the dominant TEM-like mode on the microstrip line. The pins forming the walls of the cavity are sufficiently spaced at tlhe microstrip line so that they do not disturb tile dominant mode fields. In this manner, the dominant microstrip mode can pass throughl the wall unperturbed, while the higher order modes are reflected by effectively, a perfectly conducting wail. The fields for the reflected mode, external to the cavity are then given by EK = r (e7TEM - i ETEAf) ea~B (6.2) Hi = -r (ITEM -I HTEA) eJl (6.3) where e and h are the field components transverse to the propagation direction, i. Rearranging Equation (6.1) we find thle reflection coefficient to be I HTEM l; dS r = _ ot (6.4) Il1ig (ETE: M X')TEAM) (-i) dS w~aveguade This is the central equation needed to produce the equivalent impedance of the slot from which the equivalent scattering parameters caii be derived. In order to evaluate this expression, we note that field components from the solutions to the two problems illustrated in Figure 6.2 are required. The first case - which wiU be referred to as'the waveguide problem' - is a shielded microstrip line with layers matching the cavity structure, and provides the transverse components of the field on the entire cross-section. From these we can calculate the reaction integral in the denominator of Equation (6.4) which, for the lossless case, is twice the characteristic impedance of the line. Additionally, we will have the HiTEAf term in the numerator at the location of the slot which is used as thie excitation for the second case;'the cavity problem'. In this case we need to find K, a conductor-backed, equivalent magnetic current, which replaces the tangential electric field in the slot through the Equivalence Principle [33].

132 W ith assumptions on thle symmetry of the field in thle slot as in [G67, the cavity-backed slot can be treated as a two-port series impedance element. Once r is known, we can determine thle equivalent slot impedance (normalized to the microstrip line impedance); 2r s = 2(6.5) 1-F from whiclh thle equivalent circuit scattering parameters can be derived by standard network theory. The integral equation formulations for these problems and the solutions for the currents by means of the method of moments has been described in the previous chapters. We thierefore can proceed to the verification of the theory and numerical model by presenting the experimental results. 6.2 Numerical Results and Measurements The solutions for the waveguide problem are obtained as outlined in Chapter IV. Thle cavity problem is solved using conventional Method of Moments techniques, as outlined in Chapter 111 with the HTEM field from the waveguide problem as the incident field on the slot. Both the strip and the slot are assumed to be sufficiently narrow so that the transverse components of current can be neglected. The longitudinal components on the strip are expanded in terms of piecewise sinusoidal basis functions with the Maxwellian transverse distribution as used for the waveguide problem. Galerkin's method is used to enforce the boundary conditions on the strip. For the slot, we have investigated the use of both the Maxwellian and uniform (rooftop) transverse dependence, using Galerkii's method for the rooftop functions and point matching for the MaxweUian case. In the latter case, point matching is required due to numerical considerations in the evaluation of the Sommerfeld integrals for the half-space. This requires additional computation since the matrix loses its symmetry. We would expect the Maxwellian results for narrow slots to be more accurate since the edge conditions are satisfied, however, in the end the differences

133 1.2; Shavit's Measuruneu. 1.0 c:......... Th, Te,oy /, ~r! C --- Shavit's Theory'~ 0.8 - Cosine Assumpuon u 0.6. -"~ 0.4 0.2 0.0' _ -.200 -.100 0.000 0.100 0.200 Slot Offset (inches) Figure 6.3: Normalized resistance as a function of slot offset compared to Shavit's measurements [67]. are minimal. Therefore, the rooftop functions are preferred and will be used throlighout. with the added benefit of their greater flexibility in modelling wider or more complex slots. With this discretization of the integral equations, the resulting system of linear equations is solved for the unknown current distributions by matrix inversion. To verify the accuracy of the model, we first compare with the theoretical results and measurements of Shavit [67] shown in Figures 6.3 and 6.4. Figure 6.4 shows-the results of calculations of resonant length which is a key parameter in the design of high performance antenna arrays. This is a special case involving stripline and no dielectric cover and shows significantly improved agreement between this work and the experimental measurements. The reason for the larger discrepancy in Shavit's theory is uncertain, since there are some similarities with the present method. The most likely factor may be his use of approximate Green's functions, which also restricts application of his method to striplirre only. Tllis

134 2i. I - Lo - I' I'' 2.1 2.0.. e.,} ~_ —--- Shavit's Measurnmenus C. 1.9 —........ Th eory'i ----- Shavit's Theory _1.81.7 -.200 -. 100 0.000 0.100 0.200 Slot Offset (inches) Figure 6.4: Resonant length as a function of slot offset compared to measurements [61]. leads to certain assumptions about the enforcement of the boundary conditions on the strip, whereas we have used the exact expressions and enforced the boundary condition explicitly. The number of modes used in the summations (convergence) can also be an important factor. High mode numbers are needed, especially since the critical field behavior occurs in the region where the strip and slot are most closely spaced. Since making accurate measurements for these types of slots is quite difficult and considering that these measurements were made almost a decade ago, we have performed extensive measurements on a new set of slots, with and without dielectric covers. In the process, we have uncovered new factors not previously recognized which can significantly influence the results. Individual test pieces for several combinations of slot lengths and offsets have been built. The pieces were constructed using two copper-clad dielectric boards (31 mil thick 2 inch square Duroid.5880 (, = 2.2 with 0.7 mil thick copper-cladding). The

135 size of the boards is small enough to minimize material and fixsturing costs but large enouali to provide an adequate ground plane and reduce the effects of reflections from tlhe board edges. The slots and transmission lines for each sample were produced by removing copper from the top planes of the top and bottom boards, respectively. The slot length (LS) and offset of the transmission line from the end of the slot (s) were varied from circuit to circuit with the slot always in the center of the board. The top and bottom boards were then bonded together by placing 3M 6700 2 mil thick thermoplastic bonding film at their interface and subjecting them to the heat and pressure profiles specified by the manufacturer. In addition, to improve accuracy, calibration standards were constructed for de-embedding of the fixtures using the TRL method [34]. Dielectric covers were made from 125 mil Duroid (c, = 2:2) bonded to the circuits with Ablebond 550 thermoset in a bonding press at 230~F for approximately 3 hours. Recessed tracks were milled in the covers to accommodate tlhe soldered shorting pins which protrude slightly above the top surface of the circuits. Other dimensions were held constant to the extent possible throughout the experiment and were: a = 0.425, b = 0.516, W = 0.0279 and Ws = 0.0315 inches. A sample slot is shown in Figure 6.5 at various stages of fabrication. Once calibration is established which places the reference plane at the center of tlhe slot, the S-parameters were measured for each slot. The equivalent impedance of the slot as a series element can be determined from Z = 2Z - (6.6) or z =2ZoSu (6.7) 1- Sil The extent to which Equations (6.6) and (6.7) agree provides a measure of the accuracy of calibration. In general, however, Equation (6.7) was found to be more accurate for resonant lengths because the calibration of S2l is more sensitive to dielectric constant and length

13G ~ - t, C3.r: ~ jt ~~ ~ ~ te n1~t ~t~ ~c —~ u;LVfi2k. Ctcy: ~r;f5 L. ~-i'"` ~-~ii ~-''i -- ~~'?1'~:;~~.L' Ltf -.5 J:~ ~~2*.. X'"...... ~ +:,S ~ L-4.,... z-I~ ^ r.ic~' cl`~ C rr ~t, -- ~,r ~ "' i' * 3 -r,..;Clt, ~).IAICe~.TF(IIPUI~.; \,~ r~ ~~ " \~'1* B Y.,`"LL' r " * tre J ~F(::.' ?*zi7 Z: h' qu\. ~\~r Y?::~ h, rt:l rrs ~, L- " ~,S.C.~ ~~ cc'`'f-,"~ L~* i iL "~ ii id r,.5j;.,CI.1.:t, j/ J " t. it e rr.. ".L... r 3 r:,-~c:tt,, i, ~ ~ rI`~,i~-..+ A~.> i' rY;y; r. C ~~~ Tt. a,I~a.~` \*iji:rttr.IC C~1~''~ ~Lj-. i r.'"a'.- ir~ 1 r LT r ~Q ~C,r c I C ~~~~'~~~." i kAr. *~ ~..;%-~.-~~ ~I't, u~ ~3, c. E, t. ~r -- ~-~~ Figure 6.5. Photograph of slot assembly pieces in various stages of Ibr ication.

137 variations. In contrast, Equation (6.6) was judged to be more reliable for estimations of tlie peak resistance values based on the behavior of S11 with frequency in the few cases where the results of Equations (6.6) and (6.7) differed. We have also defined resonant length as corresponding to the peak in the resistance curve, since this definition is less sensitive to measurement and calibration errors and is more consistent when constant parasitic reactances are present. Figure 6.6 shows an example of a comparison between the calculated resistance at 3.00 * - L,s a305 Mn - model 2.50 * ----- L=320in. -model. —. — L=. 335 in. - model 31.50 L.335 -in d ~ / Ic z 1.00. * 0.50 0.00 0.000 0.010 0.020 0.030 0.040 0.050 0.060 0.070 0.080 0.090 0.100 Slot Offset (inches) Figure 6.6: Uncompensated peak normalized resistance as a function of slot offset compared to measurements for uncovered slots. resonance and the measured values, normalized to the impedance of the line. The curves represent the resistance versus offset, the distance from the end of the slot to the centerline of the strip, at the measured resonant frequencies. The data points shown are the measured values with error bars indicating the uncertainty due to a combination of repeatability tests

13S and an estimate of potential sstematic calibration error due to variation in the substr.atfs By comparing the measured resistance values with the corresponding points predicted bs the numerical model, it can be seen that virtually all predicted resistance values are low. Furthermore, it should be noted that the error bars are conservative. sinice they do not account for slot and dielectric cover losses whiicil cannot be calibrated out, or variations in fabrication such as dimension changes. Also, the plotted offsets have been artificially staggered ~1 mil for clarity and do not indicate an error estimate. Fabrication variations are blamed for points which deviate from the trend of the data, for example in this Figure. the case where s = 0.075 and Ls = 0.305 inchles is particularly suspect. Figure 6.7 displays the predicted resonant lengths for the same frequenc! range. Hlere the crosses indicate the physical parameters for the measured slots. Oiie can then estimate the error in either resonant length or frequency by correlating the nmeasured resonant frequencies listed in the table with the curves. A comparison of thile resonant lengths shows that the predicted lengths are all too long. (Equivalently, predicted resollant frequencies for a given slot length are too high.) These comparisons, together with other observations made in making the measurements and numerical parametric studies, led to the important realization that a compensation must be made for the aperture in the wall which allows the strip to pass through. In the past it has been assumed that if the pins are sufficiently far from the strip, the dominant mode can pass through the wall unperturbed. At the same time, it has been assumed that if the pin spacing is sufficiently dense, the higher order modes - particularly the TElo mode - will effectively encounter a perfectly conducting wall allowing us to use the closed cavity Green's function to model the walls. Although the first assumption is valid, the second may not be and is of crucial importance because the behavior of the slot is strongly influenced by the TElo mode, leading to the above discrepancies.

139 0.370 -'.. i. I.. l'...'. i. 0.360 133GHz'~ 0.350 ) 0 3 2 0t- - 13.5 GHz -- 0.340...... -- 13.6 Gilz e~a ~~O.'~ —.'-.........'[~~~~ 13.7 GHz 0.330' - - -.......... MEASURED RESONANT FREQU ENCIES..... 13.8 GHz CZ 0.320 t3.9 G.. (VA s =4.6 14.17 — 1.014.0 GHz 0.310 0.300 1=- 0 - 4.2 G1tz 0.290 0.000 0.010 0.020 0.030 0.040 0.050 0.060 0.070 0.080 0.090 0.100 Slot Offset (inches) MEASURED RESONANT FREQUENCIES (GHz) Slot Offset Slot Length (inches) (inches) L, =.305 L. =.320 L, =.335 s =.030 14.165 13.74 13.39 s =.045 14.145 13.845 13.38 s =.060 14.17 13.80 13.31 s =.075 14.045 13.61 13.285 Figure 6.7: Predicted resonant length as a function of slot offset for uncovered slots. Crosses indicate actual test piece dimensions to be compared with the frequency curve corresponding to the measured resonant frequencies listed in the table.

140 The sensitivity of the slot's behlavior to thle relative position of thle walls parallel to tile slot is demonstrated in Figure 6.8. We see that the resonant slot length for the uncovered 0.330.....',) 0.320 c= 0.310 0 X 0.300 _ Covered, s =.060, f = 12.0 GHz No cover, s =.060, f = 13.8 GHz 0.290 0.400 0.410 0.420 0.430 0.440 0.450 0.460 0.470'a' Dimension (inches) Figure 6.8: Sensitivity of resonant length to cavity length dimension'a'. case depends on the position of these walls on a nearly one-to-one basis. Thus, a 1% error in the position of the walls will produce approximately a 1% error in the resonant length. Assuming that our average fabrication tolerances are better than that, implying that we should emphasize the trend of the data, the discrepancy between theory and experiment can be traced to ignoring the apertures in the walls for the TElo mode. (Note, however, that just the diameter of the wire used to construct the walls of the cavity - 10 mils - is on the order of 1.5% of the electrical length of the cavity.) It is observed that the covered case is less sensitive to this dimension because the magnitude of the slot field in the covered case is not as large, implying a smaller excitation of the TEio mode relative to the other

141 mechanisms. In a rectangular waveguide. wall apertures of the type used here can be represented by a shunt inductance. For a one-port, a shunt inductance can be replaced by an equivalent length of shorted transmission line. Since for stripline and rectangular waveguide the TEIo modes are identical, variatiouial formulas for calculating this inductance in rectangular waveguide such as found in 1[4. pp. 165] can be used to estimate a correction to the length of the cavity, compensating for the affect of the aperture. It is also desirable to suppress the propagation of energy in this mode thirough the aperture, by lining the stripline with shorting pins to cut off all hiiglier order mnodes outside the cavity as shown in Figure 6.5. In addition, by modelling the wires explicitly after a manner similar to 143] as discussed in Chapter IV, and using tlle'rule-of-thlumb' from [54, pp.188-189] to account for tlhe chiange in width, we have found that the correction may be somewhat smaller than indicated by variational formulas and has less sensitivity to frequency. The end result, in this case, is an estimated correction of 12.5 mils for the length of the cavity. The other walls of the cavity are also made with wires but the slot characteristics are quite insensitive to this dimension. The model is now re-evaluated with the compensated dimension and the predicted resistance and resonant lengthls are compared to the measured resistance and actual slot lengths at the measured resonant frequencies. In Shavit's case, the dimensions used result in only a 16 mil correction which, for his operating frequency, has minimal influence on the results shown previously. For our measurements the compensated coupling (resistance) results are shown in Figure 6.9 and the predicted resonant lengths are listed in Table 6.1. Parametric curves for resonant lengths for this and all remaining cases are similar to Figure 6.7 except the family of curves are shifted downward and therefore will not be shown. When the dielectric cover is introduced the same general features are observed. Using the same length compensation as in the uncovered cases because the cavities are the same, the comparison

142 3.00 — 1-....I...................... I ——. I. @-. - L, =.305 in. - model 2.50 -- L, =.320 in. - model _~ t —, —-, =.335 in. - model v) 2.00 _ i L, =.305 in. - measured L, = 320 in.- meaursed: 1.50i- L=.335in. - measured,' 1.50 -,, z 1.00 0.50 0.(0.000 0.010 0.020 0.030 0.040 0.050 0.060 0.070 0.080 0.090 0.100 Slot Offset (inches) Figure 6.9: Compensated peak normalized resistance as a function of slot offset compared to measurements for uncovered slots. Slot Length (inches) Slot Offset L,(actual) =.305 L,(actual) =.320 L,(actual) =.335 (inches) f,.. L. (model) f,e. L, (model) f,., L, (model) s =.030 14.165.304.5 (-0.2%) 13.74.3234 (+1.1%) 13.39.3409 (+1.8%) s =.045 14.145.3009 (-1.3%) 13.845.3139 (-1.9%) 13.38.3361 (+0.3%) s =.060 14.17.2957 (-3.0%) 13.80.3120 (-2.5%) 13.31.3351 (+0.0%) s =.075 14.045.2976 (-2.4%) 13.61.3175 (-0.8%) 13.285.3331 (-0.6%) Table 6.1: Measured resonant frequencies (GHz) and predicted resonant slot lengths for uncovered slots with compensated cavity length. (Deviation from actual slot length listed in parenthesis in percent.)

143 between theory and experiment for the covered cases can be found il Fiaure t 10 aild 2.00.... -...-... ~.... r.. i......I. - L, =.305 in. - model o ---. —- L =.320 in. - model - 1.50 L ---- L, =.335 in. - model._'i Ls =.305 in. - measured C: ~ L, =.320 in. - measured 0.000 0.010 0.020 0.030 0.040 0.050 0.060 0.070 0.080 0.090 0.100 Slot Offset (inches) Figure 6.10: Compensated peak normalized resistance as a function of slot offset compared to measurements for covered slots. Table 6.2. The variation of the errors in these tables is attributed primarily to variations in fabrication. It can be seen that the average error for the uncovered slots is on the order of -0.8% which is considered to be excellent. For the covered cases, the average error is approximately 3.3%, however, in this case we would etpect an increased uncertainty due to additional factors including: eariations in cover tielectric constants and bonding; a potential increase in the influence of the finite ground plane due to the introduction of surface waves; and the affect of the milled recesses in the covers, altloughl very shallow, which are not included in the model. The latter two factors are of particular interest considering the sensitivity of the resonant length to the position of the internal walls of the cavity parallel

144 Slot Length (inches) Slot Offset L,(actual) =.305 L,(actual) =.320 L,(actual) =.335 (inches) f,|,, L, (model) free L, (model) fre, L, (model) s =.030 12.40.3123 (+2.4%) 12.00.3318 (+3.7%) 11.75.3453 (+3.1%) s =.045 12.35.3125 (+2.4%) 12.03.3274 (+2.3%) 11.61.3489 (+4.1%) s =.060 12.35.3107 (+1.9%) 11.83.3353 (+4.8%) 11.56.3493 (+4.3%) s =.075 12.35.3092 (+1.4%) 11.81.3348 (+4.6%) 11.54.3487 (+4.1%) Table 6.2: Measured resonant frequencies (GHz) and predicted resonant slot lengths for covered slots withl compensated cavity length. (Deviation from tihe actual slot lengthl listed in parenthlesis in percent.) to the slot. It slhould also be noted that because the resonant length of the slot is controlled by factors wlhich do not scale identically with frequency, these errors do not indicate the expected error of predicted resonant frequencies. In fact, for the geometry studied here, the expected error in resonant frequency for a given slot length, is roughly half of the prediction error for resonatlt lengthls. This feature can be deduced from Figure 6.7 for the uncovered slots, and similar parametric plots for the covered case, by calculating the slope of length over frequency. The case for resonant lengths has been shown in the tables, since these errors are more relevant to initial antenna designs. Finally, we point out that the fixtures were built by varying the position of the strip to achieve different slot offsets while in the numerical model it is more efficient to vary the slot position relative to the strip. A comparison of numerical results for the actual dimensions has shown that the relative positions of the strip and slot in the cavity have some influence on the slot characteristics, however, the variations are within the bounds of the experiment. indicating the true accuracy of the approach is at least as good as suggested above. In fact

145 to some extent there is coupling between errors in frequency and resistance values so that variation of a parameter tends to improve the agreement withl experiment for one while degrading the comparison for the other. For example, in the covered case, evaluating the numerical model at the measured resonant frequency tends to reduce the error in predicted resonant length while increasing the discrepancy in resistance values. For the uncovered slots, the resonant length (frequency) and resistance values are more stable, suggesting not only that the compensation length is likely to be correct, but also that the introduction of the covers hlas introduced an error in the experiment. Nevertheless, the overall agreement with the experimental data is considered to be excellent to the extent that the variations can be controlled for this type of construction. 6.3 Summary In summary, we have presented a full-wave integral equation approach for the analysis of strip-fed cavity-backed slots. The formulation involves the application of the Reciprocitv Theorem which requires the field solutions to two distinct situations; the infinite shielded line (the'waveguide problem') and the cavity problem. Methods for deriving the exact Green's functions for these problems have been described including the incorporation of multi-layered substrate and superstrate capability. Solutions to the integral equations are then found by the Method of Moments. Modelling of the strip itself, and accounting for the effect of the openings in the cavity walls, were found to be a key factors not previously recognized for accurate modelling of these types of strip-fed slots. Comparison with measurements for the special case of stripline has demonstrated that solutions using the exact Green's functions and compensation for the strip aperture lead to significant improvements in accuracy over previous methods. The analysis described is also exact for more general structures involving multipie layered substrates and superstrates with the exception of determining the compensatin

146 length for the feed-through aperture. Thlis can be found by extension of tile'waveguide problem' approach as described above with wires or apertures modelled in a manner similar to [43] or [3]. For the type of fabrication techniques described in this experiment, it was found that the numerical model produces accuracies on the order of that for the experiments. This suggests the method has sufficient accuracy to replace many of the empirical steps used in the design of antenna arrays, which can significantly reduce design and development costs.

CHAPTER VII MODELLING OF THICK-SLOTS It has been suggested that strip-fed radiating slots may play ani important role in the recent trend towards more complex antenna systems. This is particularly true for monolithic phased array applications where active devices are integrated with the radiating elements for phase control and in some cases, to generate power. One of the practical difficulties which arises with the introduction of active devices is the dissipation of heat. One solution is to increase the thickness of the antenna face for use as a heat sink, which also adds the possibility of cooling channels. This has an influence on the electrical design, however, since the slots may no longer be considered infinitely thin as is commonly done. On the other hand, we can take advantage of the thickness to improve bandwidth, which has traditionally been a fundamental limitation of slot arrays. This possibility is the primary topic for this chapter. While semi-empirical and approximate correction factors for finite thickness slots have been introduced and may be adequate and appropriate in some circumstances [391, it is desirable to develop exact methods to improve the accuracy and generality of numerical design tools. The full-wave integral equation technique whichl has been presented is also applicable to this class of antennas, allowing not only the analysis of thick slots 184, 85), but also slots which couple through a section of waveguide as shown in Figure 7.1. This latter case also provides the mechanism for improved performance. By adjusting the slots 147

148 on the enids of the waveguide section to different lengths, we can essentially'stager turine each slot to a different frequency, increasing the overall bandwidth. Radiating Slot Coupling Slot Rectangular Stipline/ Waveguide Figure 7.1: Stripline excitation of waveguide-coupled slots. 7.1 Numerical Results for Waveguide Coupled Slots To illustrate the behavior of the structure, a particular case at 9.7 GHz is presented with a 0.60 x 0.60 inch cavity (c, = 2.2) fed by stripline with a ground spacing of 0.125 inches. A strip width of 0.1 inches was assumed which results in a SOf line and the slot is of the same widthl and is assumed to be covered by an E, = 2.2, 0.0625 inches thick dielectric sheet. The coupling to the slot can be represented by an equivalent series impedance. As will be seen, its value is controlled primarily by the length of the waveguide section (Lw) and thie slot offset (s), the distance between the end of the slot and the centerline of tile strip. Note that the waveguide must be dimensioned above cutoff, if substantial lengths are to be used. Thus, we have assumed half-height X-band waveguide with transverse dimensions of 0.900 x 0.225 inches. Figures 7.2 and 7.3 indicate typical impedance behaviors as functions of the slot length. Figure 7.2 shows the influence of the slot offset for the case of a conventional'thin' slot as treated in Chapter VI which also applies to the'thick' slot case. Figure 7.3 demonstrates

149 Normalized Resistance R/Zo Normalized Reactance X/Zo 1.50 1.35 0.90. 0..3 a is 1.35 7......... t 09.6 maus 1.20 0.70 1.o05 / ~ ~ ~ ~ 5 [ -''-....: =3.. 0.90 / 00..0. -'- a3 =.9,au 0.75 ~ 0.60 O. 10 "'~-~.....' = 39.4 ru. 0.4.-:..s.s.s.s~. s a 47.9 =i, ~0.31*0 0 "* 5 649ams 0.30 0.15, 73-714 sai i 0.00.050 0.350 0.375 0.400 0.425 0.450 0.350 0.375 0.400 0.425 0.450 Slot Length (inches) Slot Length (inches) Figure 7.2: Typical variation of impedance as a function of slot length for a conventional'thin' slot with slot offset as a parameter. Normalized Resistance RiZ= Normalized Reactance X/Z0 0.30. 0.40 0.25 - 035....... as 43 Ii0 0.30....3 wad 0.20 0.25 0....'I~~~~~~~.....6.lJ & a.213.3) 0.10 o0.10 P*~~~~~~~~~~~D~~...*..&- 1 m 0.10....~. iads 0.05.. 0.0.0.05 0.00 0.00 0350 0.375 0.400 0.425 0.450 0.7S 0.4 00 0:350 0.375 0.400 0.425 0.450 0.475 0.5f0 Slot LImgth (inc) Slok tLagth (indic ) Figure 7.3: Variation of impedance as a function of slot length with the difference in top and bottom slot lengths as a parameter.

150 Peak Normalized Resistance Resonant Slot Length (inches) 1.75 -. - 5, - 0..47 i.500 0.450 1.25 0.425........ 1.00 0.400...... -.-.- *-.3 ~ - 3 ~ -... L,,,.5,m 0.750 0.375 0.500 0.350'- ~.7 i 0.250 o.. 0.325 0.000 0.300 0.000 0.020 0.040 0.060 0.060 0.100 0.000 0.020 0.040 0.060 0.060 0.100 Slot Offset (inches ) Slo Offs ( inchcs ) Figure 7.4: Slot coupling (resistance) and resonant length dependence on offset and waveguide length (Lw). similar behavior for the'thick' slot case when the difference in length between the top and bottom slots (A) is used as a parameter and the slot offset fixed. Notably, the reactive part does not necessarily cross through zero. This is due to parasitic reactive discontinuities in the geometry which are not strong functions of slot length; for example, the step discontinuities in the slot-to-waveguide transitions. Nevertheless, a resonant behavior is clearly indicated and we will define resonant length by the peak in the resistance curve. We also see an affect on resonant length which is typical. The influence of slot offset and waveguide length are summarized in Figure 7.4. In this case the top and bottom slots are identical in length. Frequency response can be inferred from the impedance behavior with respect to slot length, through its relationship to wavelength. Therefore, we can use this parameter as an indication of apparent bandwidth. Using length as a parameter is preferred because this data is more easily obtained than direct frequency data. Returning to Figure 7.3 we see that as the difference in length (A) increases with slot offset held constant, the

151 1.00 0.90 0.0 0.70 t o 0.60 0.50 0 0 0.40. 0.30 0.20 o 0 a 3L3 mU. s = 11.9 mis Z/0.10 -— 0 —- & wO.Omils. s-39.4ma k 0.00 0.90 0.95 1.00 1.05 1.10 1.15 1.20 Normalized Slot Length L, / L,,5 Figure 7.5: Apparent bandwidth comparison for identical slots versus slots with different lengths based on an interpretation of the relationship of slot length to wavelength. apparent bandwidth is increased together with a change in resonant length, but the coupling resistance is simultaneously reduced. Since the resistance is normally a fixed value dictated by the feed design, we can compensate for this either by increasing the slot offset or adjusting the waveguide length as indicated by Figure 7.4. However, because apparent bandwidth reduces in a similar way with increased offset, it is difficult to judge whether any gains in bandwidth can be produced. To evaluate the improvement in bandwidth for the largest slot length difference of Figure 7.3, the offset was varied wlhile the waveguide length was fixed at Lw = 0.5 inches. The normalized resistance curves were then re-normalized to the peak value to remove any remaining discrepancy (R/Zo = 0.294 vs. 0.295). Slot lengths were also normalized to the resonant lengths to remove variations. A suggestion of an increase in bandwidth is now obvious as illustrated in Figure 7.5. The apparent bandwidth interpretations so far have been based on the slot length. As

152 0.35.,,, 0.30 0.25 0.20 ~ 0.15 0.10 z —. - s.1.9. uL< -.63. 1,.,, -,~ e, ) 0.05..... s * 39.4. L_ -1.. * 417 7 (ms) 0.00.... I... I.... t I..... I..... I,..- 9.4 9.5 9.6 9.7 9.8 9.9 10.0 10.1 10.2 Frequency (GHz) Figure 7.6: Apparent bandwidth comparison for identical slots versus slots with different lengths based on calculations at selected frequencies. indicated earlier, however, each transition in the structure may have parasitics impedances which are independent of slot length. These are unlikely to be constant or have the same variation with frequency, therefore, to rigorously translate the previous figures into a frequency response, it is necessary to run each geometry at an ensemble of frequencies. The plot in Figure 7.6 shows a comparison of the two cases over a sample of frequencies. It is evident that the previous estimates based on length calculations were conservative, so that we may be justified in a preference for the length-based results, at least for a parametric study of the structure. Comparing Figure 7.6 with Figure 7.5 also suggests that although, strictly speaking, scaling cannot be applied to the length data to derive the frequency response, it does provide a reasonable estimation of the actual response, at least in this case. Scaling length data could then be used in the initial design stages to increase the efficiency of the design process.

153 7.2 Summary and Conclusions In conclusion. the thick slot hats been modelled by treating the slot apertures as irc-i rate slots coupled through a section of rectangular waveguide. It has been show n through a parametric study that, based on the theoretical model developed, the apparent bandwidth of the element can be enhanced by stagger-tuning the coupled slots. Although this part of the analysis has not been verified by experimental evidence, the concept is a simple exstension of the models verified previously and the behavior of the structure follows intuitive expectations. Within the limited range of parameter variations used here, apparent bandwidthl increases of 50% are possible. The true bandwidth would depend on the external circuits actually used to feed and load the slot. It seems likely that with a more detailed study of the influence of various parameters on the impedance characteristics, even greater improvements in bandwidthl could be obtaiiied.

CHAPTER VIII CONCLUSIONS AND RECOMMENDATIONS FOR FUTURE WORK The class of structures treated in this work involves rectangularly shielded microstrip and slot lines. An integral equation formulation for the analysis of these structures in various combinations has been shown to provide accurate characterizations of a variety of elements. The capability to analyze multi-layered substrates and superstrates has been included throughi the use of impedance boundary conditions and provides an exact representation by modal analysis of the geometries involved. Solutions to the integral equations require the specification of Green's functions for each case. A number of Green's functions of different types have been derived by a extending and generalizing a dyadic analysis technique to incorporate the impedance boundary conditions. This approach greatly simplifies the treatment of multilayered structures through the use of transmission line analysis. The impedance boundary conditions have thlus been incorporated into the exact dyadic Green's functions making the analysis "full-wave", therefore all electromagnetic interactions are taken into account. For all structures treated, the derivation of the Green's functions reduces to a straightforward procedure producing all components of both the electric and magnetic field dyads in one exercise. Because of the normalization of the functions used for the expansions of the fields, the approach also produces the dual Green's functions by a simple change in notation, similar to the application 154

155 of the Duality Principle. The integral equations have been solved through a conventional application of the Mlethod of Moments. The treatment of basis functions in general coordinate systems has been sliown to reduce to straightforward coordinate transformations. This reduces the required integrations over source and weight functions to simple forms which have been evaluated for piecewise sinusoidal basis functions with both constant and Maxwellian transverse dependence. These results lay a foundation for the treatment of coupling between strips and slots with arbitrary planar orientations within a rectangular cavity. The remainder of the work is a study of special cases of current interest. Overall, the method has been shown to produce numerical models with excellent accuracy such that one could expect to design devices involving these elements with little or no empirical work or adjustments. This conclusion is evidenced by extensive experimental data provided throughout. The remainder of this chapter is devoted to discussion of specific findings for the various applications studied and recommendations for further development of these techniques. 8.1 Summary of Achievements 8.1.1 Two-Dimensional Analysis A number of geometries have been considered in Chapter IV which can be treated by a two-dimensional analysis. The primary motivation for the studies in Chapter IV was to support the work in later chapters, particularly Chapter VI. However, the results of thlis chapter have some significance themselves. The benefits of using LSE and LSM modes for the expansion of fields in connection with a transmission line representation of impedance boundary conditions was demonstrated. It has been shown that in order to find the propagation constants of microstrip or slot lines.

156 we first need only consider the boundary conditions in the region of the strip or slot. The remaining lavers are represented by the impedance conditions, simplifying the numerical implementation. Once the propagation constant for a particular mode is known, the fields in the remaining layers, and hence the characteristic impedance of the line. can be found by a summation of modes with coefficients specified by analytically matching the modes across layer interfaces. The matching procedure results in simple expressions as a result of the use of the LSE and LSM modes. The fields throughout all layers can be efficiently found in the same manner, allowing a visual representation of the field structure which is useful for intuitive as well a quantitative information. Comparisons with available data were made to verify reliability. The presentation also serves as an illustration of the technique when applied to otlier structures such as cavities, parallel plate waveguides, or open structures. We also developed a simple technique for the evaluation of scattering from vertical wires or pins in a rectangular waveguide. The metlhod is a simplification of more complex routines discussed in the literature, allowable in our case since we are concerned with wires of small diameter. This work was needed in support of Chapter VI but also led to certain observations which have not been widely discussed. The validity of the results was verified by experimental results and published data. The specialization of this problem to homogeneously filled rectangular waveguides allows it to be treated as a two dimensional problem which precludes its use with multilayered structures; however, the extension to the three dimensions is not difficult. 8.1.2 Couplers Chapter V contains a study of coupling from microstrip to microstrip through a narrow slot aperture. The unique feature of the treatment of this problem is the use of the Standing Wave method with even and odd excitation analysis to extract the circuit parameters. This

157 method has been used by others and thus is only briefly discussed, however, some of its kev limitations are pointed out. The primary significance of this topic is in the numerical and experimental results. It has been shown that this structure enables one to make very wide band transitions from one line to the other. Thle particular structure studied, a transition from one substrate level to another, has significant implications for vertical interconnects in monolithic circuits since no via-hole is required. Qualitative explanations for the behavior of the structure, as a function of various parameters, were provided in connection with an equivalent circuit to develop an intuitive understanding of the device. The numerical analysis has been shown to produce highly accurate results by comparison with experimental data. 8.1.3 Radiating Slots The modelling of microstrip-fed radiating slots is one of the principle topics of this work. The fundamental results for this element are discussed in Chapter VI. The limitations of the Standing Wave method developed for the coupler require a different technique to derive circuit parameters. An alternative method, applied earlier by Shavit 167), was developed for our use and is here referred to as the'Reaction Method'. The technique is based on the Reciprocity Theorem and is applied in a manner similar to that used for waveguide-fed slots developed many years ago. However, due to the presence of the strip, the method is somewhat more complex and required the development of techniques described in Chapter IV. The explicit treatment of the strip for this type of antenna element, was shown to be the likely cause for discrepancies found in earlier work. Here it has been shown that the numerical model is capable of producing accuracies on the order of experimental work. The model demonstrates much better performance than earlier work on this type of slot. One of the key discoveries in this area is that the

158 openings in tile cavity wall whlich allow access for the strip cannot necessarily be ignore(d a has previously been assumed. A technique has been developed, supported by the modei> discussed in Chapter IV, which compensates for the effect of the opening. The resulting data shows excellent agreement with an extensive experiment. 8.1.4 Modelling of Thick Slots The motivation for making a thick slot may be for structural purposes or to act as a heat sink for active devices. The original formulation in Chapter I outlines the modelling of thlick slots by replacing them with a closed cavity and equivalent magnetic currents on the slots. Thus, the cavity dimensions do no have to correspond to the dimensions of the slot, allowing greater flexibility in applying the model to different applications. In Chapter VII, we numerically investigate the behavior of a structure in which the strip-fed slot couples to the radiating slot through an intervening section of rectangullar waveguide, dimensioned to allow the dominant waveguide mode to propagate. The most significant feature of this chapter is the study of the effect of allowing the strip-fed slot and radiating slot to have different lengths. This detunes the structure by stagger tuning each slot to a slightly different frequency, resulting in an apparent increase in bandwidth. In order to achieve a constant slot impedance, the strip offset and/or waveguide length must be adjusted simultaneously with the alteration of slot lengths to compensate for a change in resonant resistance. This is significant, since bandwidth is one of the primary limiting factors for these types of antenna elements. Experimental verification of this result is left for later work. However, it is expected to be correct since it is a relatively straightforward extension of the remainder of the model which has been validated by experiment. The result is also appealing from our intuitive understanding of the slot behavior.

159 8.2 Model Limitations Aside from resource issues such as tihe matters discussed in section 3.G.3. there art several technical limitations of the methods used in this work which should be pointed out. Some have already been discussed regarding the Standing \Wave method, however. the stated limitations of this technique could potentially be circumvented. For example. it has been stated that this method requires a minimum of about one wavele.igth of line away from the discontinuities in order to estimate the guided wavelength. In some cases, this information can be derived from other methods similar to the two-dimensional analyses derived in Chapter IV. Another restriction is the requirement of only one propagating mode on the line. This constraint, however, is a result of our simple interpretation of the standing wave pattern by locating the relative positions of the peaks or nulls. It is possible. by more sophlisticated parameter estimation schemes, to derive the required informationl even when more modes are present. Of course, these variations may introduce their own requirements which may or may not be beneficial. A more serious limitation involves the two-port microstrip-fed slots discussed in Chapter VI. It has been shown that the feed-through apertures, which allow the strip access to the cavity, can have a significant influence on the behavior of the slot. While we have demonstrated that this effect can be effectively compensated for in the cases studied, numerical experiments have shown that this technique is limited to strips which are significantly offset from the cavity centerline (perhaps 25% or more). It is suspected that this is due to certain assumptions made about the behavior of the higher order modes at the feed-through apertures. Specifically, the first-order correction technique we have applied does not account for potential interactions between the wall aperture, the strip and the reactive fields of higher order modes. This interaction may have a profound influence on the amplitude and phase of the TE1o mode, as well as the dominant mode, which in turn

160 may significantly alter thle field onl thle slot. The effect would be enlianced for a centered strip since. niot only is the TElo mode more strongly excited at the slot. but also it is more strongly) coupled to the feed-through aperture. Tlhus, the field on the slot may not be correctly determined under these conditions and will lead to incorrect slot impedance values. The rectification of the situation is, of course, to expand the model to include an explicit representation of the feed-tlhrough apertures, which would fully account for their influence. Other limitations of the models are not unique to this particular approach and need not be discussed in detail. For example, all numerical models involve finite approximations to reaLitv and therefore involve some errors in their representations. Nevertheless, full-wave integral equation techniques such as used here, have been proven to provide some of the most accurate and reliable models available for the simulation of a wide variety of problems. 8.3 Recommendation for Future Work Several applications should be furtler investigated based on the formulations presented and form the basis for extensions to this work. The first instance would employ the transformations derived in Chapter III to study the case discussed in [621. This case is of interest because the sensitivity of the impedance to the strip offset for the geometry of Chapter VI may place unduly stringent tolerance requirements on the fabrications process. The application of the full-wave techniques to-this problem may make possible the elimination of the semi-empirical factors used in [62]. Since the feed-through apertures could be significantly offset from the cavity centerline, the apparent problem with their neglect in the present model may be avoided. Another case to consider is the'T-Bar' fed slot which has the geometry illustrated in Figure 8.1. The formulation for extracting the impedance of the equivalent impedance could follow the Reaction Method used in Chapter VI by applying the Reciprocity Theorem

161 Short at walls L ~'~Cross Bar Opening Detail Slot Input Figure 8.1: T-Bar fed slot geometry. to the two cases shown in Figure 8.2. Here, in both cases, the strip feed line is terminated by the cross-bar within the cavitv. We now identify the reflected fields supported by the currents on the strip in the cavity, case'b', as EK' = rS (eTEM - i ETEM) e (e. ) HAT = -rS (hTEM-i HTEM)e (8.2) These are the fields associated with the dominant stripline mode produced by the scattering of the slot as before, and it is still assumed that the dominant mode is unaffected by the required aperture in the cavity wall. The fields supported by the currents in the'a' case, however, are now given by EL = (e-in + rLe'')eTEM + (e-j - -rLe'0) l. ETEM (8.3) HL = (e-'1t- rLe.'' )TEM + (e-0ot + rLe')i. HTEM (8.4) where rL is the reflection coefficient of the T-bar line without the slot present, referenced to the location of the slot on the line, or the reference point for the slot, if not.at the plane

162 slot position _ H TEM a) E TEM HTEM - waveguide aperture slot position - _ b) K HK - waveguide aperture Figure 8.2: Two cases for application of the Reciprocity Theorem for the T-bar fed case of the slot. The reflection coefficient on the line is now given by 1 ||l TEM H K dS rS= - - - rL (8.5) Jufatgd( ETEM X HTEM) ( —I) dS waveJucide We now also need to know rL from which we can find HTEM which we then use as an excitation for the cavity problem to find K, as before. Accurately knowing all these terms, we should be able to produce the reflection coefficient for the slot from which the slot impedance and circuit parameters can be derived. One difficulty in this approach is forseen to be the determination of rL. This problem would be straightforward except for the fact that for the dimensions of the cavity, more than one mode can propagate on tihe line. To derive rL for the dominant mode on such a line would require further development as discussed above. Also the question of the effect of the feed-through aperture needs to be resolved which by itself can be viewed as an extension of the treatment of wires In Chapter IV.

163 Another topic which should be further developed for application to the radiatng slots is the use of a power balance formulation. If we assume thiat the walls and materials in the cavity are loss-less, which we have done throughout, then thie resistive part of the impedance for the T-bar fed slot is directly proportional to the total radiated power from the slot. This can be determined by integrating the radiation pattern of the slot field over the entire half space which can be done in the far field, given the field on thle slot. For uncovered slots, this is a straightforward procedure since the Equivalence Principle can be used to replace the slot magnetic current by an equivalent magnetic dipole current in free space for which the total radiated power can easily be derived. For the covered slots, the situation is a Little more complicated, due to the presence of surface waves in the dielectric; however, the integration can be performed in the far-field, in whichi case asymptotic expressions can be used with steepest descent path integration techniques. This method has been applied to other problems in the past where the slot voltage distribution has an assumed form at resonance [21, 58, 67, 62J, but should be equally applicable with the present technique wlhere we can determine thte actual distribution at resonance. Finally, it is suggested that the model could be extended to allow arbitrarily shaped slots and stripline configurations. It is well known that a wider slot has a broader bandwidth and that loading of the slot ends with different shapes also has some influence. This capability could be achieved by using rooftop basis functions which approximate the current in a piecewise fashion in the two planar slot dimensions. As the slot is made wider, however. one would likely need to model both directions of current as described in Chapter 111.

APPENDICES 164

165 APPENDIX A Vector Wave Function Expansions and Relations By convention, L, M and N are used througliout the text to denote the Vector W'ave Functions (VWFs) defined by L = V' (A.1) M = V x i, = -V x N (A.2) 1 1= VxN = - Vxi -Vx M (A.3) These form a complete set of solutions to the homogeneous wave equation V x V x F - K2F = O when P is the scalar function solution to the equation V2 + K2'1 = 0. The particular solution for P is chosen to satisfy the boundary conditions of thle problem. The x, unit vector is called the'piloting vector' which determines the forms of L, M and N and K is the separation constant K2 = kx + ky + k. In this appendix, the expansions of various vector wave functions (VWFs) used throughout the text are provided as a convenient reference. In addition, some identities used to reduce some of the derived expressions to simpler forms are provided. By convention, L, M and N are assumed throughout to be formed from the generating function'P = e2i(kzz+kV+kaz) with z as the piloting vector unless indicated otherwise by additional subscripts, superscripts or in the text. A common superscript to be used will be a prime (') which will imply both that the z, y and z dependence will be x', y' and z' corre

166 sponding to source coordinates, and that the wavenumbers kz, k, and k: will be led by a? instead of a -j if the functional dependence is the exponential. e.g.. ei'(t+99v Also, a notational convention lhas been established which shlould eliminate ambiguity when subscripts are used to imply trigonometric dependence. Specifically, subscripts'o' and e' are used to intdicate sine and cosine variation of the generating function. respectively. When these are used, the arguments will be provided in one-to-one correspondence to indicate the variable to which the trigonometric function applies. For example, the function Meo[kz, ky, k:(z - c)] implies that the generating function is T1([k1, kj, k,(z - c)I.'eo must then have the form [cos k,z sin kyyeJkz('-c)1, since the first two arguments are ks and kA, corresponding to the'e' and'o' subscripts, respectively. This example also illustrates the additional specification of spatial coordinate dependencies as in (Z - c) instead of just Parallel Plate: (k.r) = coskzze-J(kvy+kz) ( 4) M,(kl) = V x,Pee(k,); = [-:jky cos k, + yk, sin k:z]e-i(kv+k+*) (A.5) Ne(kr) = x V x xIe(k,)iz K = 4[ijkrk, sin kzx -.kk,, cosk,zX + i(k2 + k2)cosk,ze-j'(kY+k z) (A.6)'Po(kz) = sin krze-j(i'Y+ki") (A.7) Mo(k) = V x Coo(k,); = [-ijk, sin k,z +.k, cos k:]xe-)(kvY+kt') (A.8) No(k,) = x V x o(k,)i = [ijkk, cos kz - kl;,t, sin k.x - i (kq + k) sin tz]e-Ji(i1'+k5z) (A.9)

1G67 Rectangular Waveguide:,ee(kz,ky,) = COSkk=zcoskye -k, tA. 10,,ee(k,,k,) = V x,,(kr.k,) = -i: k, cos k sin ky + ksin k,xcos ksin]e-': kA i I.1 Neer(kz ky) = -V x V x Pee(kr ky ) = -[i jk=k sin k=z cos kyy + y jkyk cos krz sin k,y + i (k~ + k ) cos kxz cos k,yle-k' (A.12) avoo(kz,ky) = sink,xsinkyye-)k,z (A.13) Moo(k=, ky) V x too(k, kV); =[i k, sin kzx cos k,y - k:: cos krz sin k,y)e-J)k (A 14) Noo(k=, ky) = xV Moo(k=, kv) = — i jk,k cos krz sin k,y - y jk,k sin krz cos k,y + (k + k2) sin krsin n k,iyle-J' (A.15)',-o(k=,k,) = cos k,z sin k,yel ks&' (A.16) Meo(kr,k,) = V x eo(k,,ky)i = [ik cos k,x cos k,y + y kx sin krz sin kiyle-ikz (A. 17) Neo(k: ky,) = -V x V x,,eo(k, ky) = -[ijkzk: sin k:z sin kvy - jjk,k, cos krx cos k,y + i(k + k,) sin k/z sin lky,e-y) (A. 18 ),o'(k=,k,) = sin kz coskyye-yCJL (A 191

168 (-ik, sill krx sin kyy - yk, cos kzx cos k.y]e-J': A.20) Noe(kr, k) = -V x Moe(k, k) = -[-ijk,kz cos k;I cos kyy + yjk,k, sin k,x sin k,y + i(k + k2) sin k, cos k,ye- (A.21) Rectangular Cavities: Meee(kr ky, kz) = V x [ cosk,z cos ky cosk,z] + -:i ky cos k,x sin k,y cos k,z + k, sin kr: cos k,y cos k, z (A.22) Meeo(kz,ky,k ) = V x [ cosokszcoskyy sin k,z] -: k, cos kt Zsin k,y silI k: + y kz sin krz cos k,y sin k.z (A.23) Mooo(kz, ky,kz) = V x [zisinkz sin k,ysii k z] = +i ky siin k,x cos kyy sin k,z - - kz cos kz sini kyy sin k,z ( A.24) Mooe(kz, ky, kz,) = V x [i sin krz sin k,y cos k,z] +i k, sin kzz cos kyy cos k,z - y k~ cos k,r: sin kyy cos k,z (A.25) Neee(, kr, k, k) = -V x k, ) = -[+i k4k, sin kxz cos kyy sin k,z + y kyk, cos k,z sin kVy sin k,z +i (k2 + k2) cos kr~ cos kAy cos k,z] (A.26) Neeo(k,k,k) = -V x Mo(kr,k, k) 1-.i k,k, sin k,z cos k,y cos kz - k,k, cos k;z sin kyy cos k2' +i (k2 + k ) cos k.z cos k,y sin k, z] (A.27) Nooo(k, k, k,) = V x Mooo(k,,,k1 kz) -[+: kzk, cos kxz sin k/y cos k,z + y k~k, sin kzx cos k/y cos k,z +z' (k + k()sin k, sin ky sin k,z] (A.28)

169 Nooe(krx,kg, kz) = iX looe(k k.kY,k:) -[-i k,k, cos kzx sin k,y sin k2, z k,k, sin k,r cos;y sin k. +z (k2 + k2) sin k,z sin k,y cosk, z] (.9 + x y (A29) Identities: e'MktC(kz) e-'~'CM(-kz) = M[kz(z - c)] ~ Nl[-k,(z - c)] +2M,[k,(z - c)] (A.30) -2jMo[kz(z - c)]'k,CM(A.:) ~ e-,kc"M(-k,) = hI[k,(z - c)l ~ M[-k.(z - c)j +2Me[(k(z - c)] (A.31) -2jMo[k,(z - c)] eJ'2CN(kl) ~ e)kIcN(-kz) = NRk(z - c)] ~ N[-k,(z - c)I +2Ne,[k.(z - c) (A.32) -2jNo[k,(z - c)] ek,zN(kz; ) e e-J'kCN(-kz) = NF[k,(z - c)] ~ N[-kz,(z - c)] +2N,[k,(z - c)] (A.33) -2jN,o[kz(z - c)] ecMe(k,:kz) ~: e-J" Me(k, -k) = M,e[k,,k.(Z - C)l ~ M,[k, -k,(z - c)] +2Mee[k:, kz(z - c)] (A.34) -2jMeo[k,, kz(z - c)] eJ,-'Mo(kx,kz) ~ e-k"CMo(k, — kz) = Mo[k,,k.(z - c)] ~ Mo[k, -kz(z - c) +2Moe[kt, k,(z - c)] (A.35) - 2jMoo[k,, k,(z - c)] eJ'.CN,(k,::kz) ~ e-.J'cNe(kx, -k,) = Ne[k,,k,(z - c)] + N[kr.-k,(z - c)I +2Nf,e[k,, k,(z - c)J (A.36) I -2jN.,[1k.,k(z - c)] +21qoe[k,, k,(z - c)] =1k.clk-T /l- 1- % I — Ik.Ck-l f 1- 1- KT f( A 3, ) t-i[_9,~'. -(. _ c,

170 eJ kCMe(k=, ky, kz) ~ e-Jk'c.ee(k=, ky, -k ) = M,[k=, k., k(z - c)] ~ Mle[k,, k,, -k:(z - c)J +2Meee[kz, ki, k:(z - c)I -2jMeeo[kr, k,, k(z - c)] ek*ci M (k, ky ) k)) i e-JikCMoo(k=, ky, -kz) = yMoo[k, k,(z - c)] ~ Moo[kr, kv, -k:(z - c ) ( +2Mooe[k,, ky,,k(z - c)] ~~~~~~~~~= < ~(A.39) -2jMooo[(k, k,, k,(z - c)] elk*c,Ne (kr, k, ) e- N e -J'C5ee.(k,k, -k,) = Neek, ky, k, (z 2- c)] ~ )Ne[kz, ky,-k z - c) +2N,,ee[kz,, ky k,(z - c)] =i~ o[t y (A.40) -2jN[eo[k, ky, k,(z - c)] eifkt'oo(k, k, ky ) I e-krcNoo(kr. ky, -k,) - Noo[k, k,, k(z- c)J ~ Roo[kx, ky, -k,( - c) +2Noo,,e[kz, k,, kz(z - c)] =t~~~~~~ ~~~(A.41) -2jNooo[k, k., k2,(z - c)]

171 APPENDIX B On the Use of Vector Potential Functions for the Derivation of Green's Functions In previous literature there has been considerable discussion on the treatment of sources in waveguides or cavities, particularly tile problem of a longitudinal slot in the broad wall of a rectangular guide. It has been stated and generally accepted that an additional "mode" needs to be included in thle Greeii's fuiictioii for tile waveguide to accurately represent tile currents at the slot [86]. However, the origin and derivation of this mode is somewhat unclear. The difficulty arises from an incomplete treatment of the potential theory in the source region for these problems. It is identical to the difficulty associated with the vector wave function or direct field expansion formulations for which the problem has been adequately resolved after much discussion in the literature over an extended period of time (81. 6.5. 42, 15, 77, 11]. Because the use of vector wave functions as used in this work is perhaps somewhat unfamiliar and since the difficulties for the vector potential method are very closely related, this appendix presents a discussion of the use of potential theory for source regions to demonstrate the origin of the required'additional' terms. B.1 Potential Theory Solutions for Electric Currents To examine the method, we will first consider the fields in a rectangular wavegul(ic excited by a point source of longitudinally directed electric current. The waveguide.xx s

172 assumed to lie along the i direction. We must find solutions to Maxwell's equations as follows: V x E=-jWpH (B.1) V x = j, + 6(z -')b(y- y')b(z- z') B.?) where the assumptions stated in section 1.3.1 are employed and in which we have introduced the current source J = 6( - z')6(y - y')6(z - z') (B.3) Since V H = 0 we can represent H by H = -V x A (B.4) where A is, as yet, an arbitrary unknown vector function. Using Equation (B.4) in (B.1) we find Vx E=-jwVx A (B.5) which we can integrate to get E = -jw A + V9 (B.6) where 4 is a scalar function representing the constant of integration. Using Equation (B.6) in (B.2) we find the relationship between A and 4 must satisfy V x V x A - k2 A - jwiV = uJ (B.7) As A is arbitrary, its specification determines 4 through Equation (B.7), to within an incidental constant. Equivalently, 4 may be arbitrarily specified which in turn defines A. Both, of course, are not entirely arbitrary since we must consider the resulting boundary conditions on the E and H fields. Thus, we are free to choose 4' in order to simplify

173 Equation (B.7) to a form for wliclh the solution is eitler known or mav be rcad! foundi This step is really the quintessential motivation for the use of thle vector potenitial 11et Ilod. since the overall problem is reduced to the solution of a sigle presumably sinple. equation. The selection of two conditions which specify.4 or 4 may be familiar. The first is the Coulomb condition, or Coulomb'gauge', in which A is assumed to satisfy V. A = 0. From Equation (B.7) it can then be shown that ~ satisfies Poisson's equation: V2 = J V.J (BA.) however, this does not lead to a simple equation for A. Tlius, tiis approach is not generally used [53]. The Lorenz gauge [82] is by far more commonly used in wlich it is assunied tlhat V A = jwju41. This reduces Equation (B.7) to (V2 = -V x V x +VV- ) v2 +k,'2 A = -SJ i B.9) To satisfy this vector equation requires that we choose A withi a com ponent in the same direction as J, reducing it to a scalar equation. In our case, for example, we set A = A. reducing Equation (B.9) to V2Az + k2A, = -Ai6( R - Rl) (B.10) the solution of which can be found by separation of variables and is the primary topic for much of the remainder of this appendix. Before proceeding however, we must pause to distinguish this approach with other approaches involving potential functions. It is widely recognized that the fields in waveguide can be expressed in terms of a variety of combinations of potential functions, most notably in this context, the A and F potentials. For example, the complete set of homogeneous solutions for rectangular waveguide can be written in terms of A = A,i and F = F'

174 where H = i x A and E = - V x F yielding the TM and TE waveguide modes respectivelv. Similarly, if we use A = A,i and F = F:i. we produce the LSM and LSE modes which are TM and TE to the z direction respectivelv. Various other combinations are possible, not restricted to the rectangular waveguide problem, as noted in [25] for thle dielectric lhalf space. We should recognize that these potentials do not proceed from an attempt to solve the inhomogeneous field equations through the use of potentials, as above, but rather are solutions to the homogeneous (source-free) equations which allow a great deal more flexibility. Solving the inhomogeneous case by expanding the fields in terms of homogeneous potentials and enforcing the boundary conditions at the source, amounts to a field expansion metlhod closely related to thie vector wave function methlod employed in the main body of this work. As sucli, they perhaps should not properly be referred to as'Potential Theory' methods as is often done. It is in these cases that the functions themselves can truly be called'auxiliary potentials', since they are not directly solved for. but rather are only a vehicle to express the field in a simple way. It is also noted that these methods often introduce the radiation condition into the potentials whiich prevents them from being truly homogeneous solutions as will become evident. Returning to the solution of Equation (B.10), the particular solution to this inhomogeneous equation can be constructed from the general homogeneous solutions. Once A, has been found, the fields will be given by H = -V x (A,i) 10A,. 1OA,z 1- z - y (B. 11) M ay A aX + VV (Azi) E = -jwA,i++ jw%A 1 ) a'A, I'Z + a + j 1w a8z - jwA (B. 12 The homogeneous solutions are readily found by separation of variables and caul 1,.

175 written in the form A: = [.4 cosk z + A2 sin k,z][B cosk,y + B2sin k,yJ[Cle+tk'-z:'); C(e-.~:(-':' (B.13) where k2 = k2 + k2 + k2. Applying the boundary condition ni x E = O at the walls of ttie waveguide reduces (B.13) to Az = sin k,z sin ky[Cle+J'tZ-z') + C2e-J't -'] (B. 14) where k, = m7r/a, k,, = n7r/b, and (m,n) are any integers. The eigenvalues k, and k, are thus infinite discrete sets while k, has a continuous spectrum. Before proceeding, it is instructional to verify that this solution satisfies the homogeneous form of Equation (B.10) which can be readily done by direct differentiation. In rectangular coordinates the V2 operator is V%2 = + v +? which allows tie x aid y partial derivatives to be performed by inspection leaving -k2 Az - k2Az + 2 + 2Az = 0 (B.15) Since k2 = k2 + k2 + kA2, Equation (B.15) reduces to O2AZ +- k2A, =O (B.16) Since A, is continuous in z, the second derivative can be performed without difficulty so it is obvious that (B.16) is satisfied and (B.14) is confirmed as a solution to the homogeneous equation. We now proceed as usual and impose the'radiation condition' on the solution; that is, E -0, H —0O, as Izl- oo (B.17) with finite losses. (Or equivalently, require outward travelling waves outside of all source regions.) Of course we have anticipated this condition in the form of the soluition's

176 dependence and we now set C'le-Jk(~-' _>' Az = sin k,xz sin k,y 1., 3 i C2eJk,(z-z') -<' Notice the implications of imposing the radiation condition. Applying the left hand side of the scalar wave equation (B.10) we can immediately see that the solution (B.18) remairis homogeneous in the entire domain with respect to the x and y dependence. Applying the remaining operations - left hand side of Equation (B.16) - we find 9AA = sink -,zsink gy kICie-k,(z z) > z' -— Z= sin k,z sin k,y + [CI - C2] 6(z -') aZ,C2eJk(z-') z < z' (B.19) Note that the term involving 6(z - Z') comes from the discontinuity in A, that may exist if Ct $ C2. We have performed the derivative by noting that for a function as shown in Figure B.1 a singularity appears at the discontinuity. Note also that the singularity may appear and be defined in the sense of a generalized function at z = z' whether or not th( original function is defined at that point [26, pp. 21-22]. We can eliminate the singularity in the first derivative by requiring C1 = C2 = C (B.20) This is a boundary condition is based on the argument that the potential must approach the same value from either direction at z = z', i.e., it is continuous [13, p. 160]. Then Equation (B.19) becomes aAz 4-e~k, ('-; ) z > ZB However, Equation (B.21) is discontinuous at z = z' and the second derivative is a2 = -k2C sin k,m sin k,y elt (z' eku(z-') z < z' -2jk,C sin kmz sin k,y 6(z -') ( B.22)

[ fo) - f(o ] 6(x + + f (x) f (x) df (x) df (x) dx cdx df+( ) df(z) d> dd( z ) - dx + [f+(x)- f-(Z)1j6( -x') dz df-(z) z < z' dx Figure B.1: Thie derivative of an arbitrary discontinuous function. Using Equation (B.22) in (B.16) we find that Equation (B.18) is indeed a homogeneous solution for A, except at the point z = z'. Notice that this result was obtained by imposing the radiation condition alone. The continuity condition was also used but effects only the nature of the singularity in the second derivative at z = z', that is, whether the delta function is produced or its first derivative. Thus, as is intuitively obvious, imposing the radiation condition dictates that the potential must satisfy an inlhomogeiieous equation with a driving function (source) located at some location between -oo < z' < oo. Notice that we arbitrarily chose the form of the z dependence so that the discontinuity would fall at z = z'. This is, of course, the most reasonable choice, since we ultimately are attempting to solve the source problem and we do not expect discontinuities in thle fields in the sourcefree regions. However, we have not yet attempted to satisfy the original nonhomogeneous equation given by Equation (B.10).

Equation (B.22) suggests that we can take (B.18) as thle homogeneous solutions for Equation (B.10) if we exclude thle poiiit z = z' from the homogeneous solution's domain We thus avoid the inhomogeneous contributioii of thte singular term in tile second derivative. The complete set of homogeneous solutions of this type (TM) are then oo e- z > _/ Az= dk. E Cm, sill k, sin k,y (B.23) - m.n < J') < The homogeneous solutions can be added to the particular solution with arbitrary coefficients without violating Equation (B.10) as long as thle the exclusion of the point =' is observed, even (especially) for derivatives of A,. Specifically, this means that the fields arising from the homogeneous solutions are valid everywhere but the z = z' point. Homogeneous solutions are often added, for example, to satisfy new boundary conditions introduced by adding obstacles to the waveguide. In fact, this is the basis for the method of scattering superposition used extensively in this work. The form of the particular solution is suggested immediately by the singularity encountered in Equation (B.22). Therefore, we assume the particular solution hlas the form e e_J k(z) z > z' At = Z C,, sin kmz sin k,y (B.24) m.nn eJkz(z-z~) z < zt Note the exclusion of z = z'. In this case, however, we do not exclude z = z' from the solution domain. As will be seen, this allows the discontinuity to contribute only when the second derivative at t = z' is evaluated and allows the particular solution to be found. This type of exclusion is allowed since this potential is a Green's function which is always defined in the sense of a generalized function. Generally, this means that the function itself has no useful meaning except in the context of an integral, in which case, non-singular isolated points can be removed without effecting the result. However, the point is an essential point at which the derivatives must be allowed to exist (in the generalized function sense) in order to satisfy the differential equation.

179 On applying Equation (B.22) then, we are left to find C,,mn such that (B 3.10) is satisfied. tilat is, -2 E Cmnk, sin km sin kny 6(z - z') = -6( -'( y - y')6( - Z') (B.25) or E cmn sin kmz sin kny = 2 - z')6(y - y') (B.26) m,n We can find C,mn by multiplying Equation (B.26) by sin kmx sin kny and integrating over the cross section of the waveguide getting Cn -j2b sinkmz' sinkny' (B.27) k,ab Equation (B.26) thus implies 4. sin kmzsin kn,Ysinkmz' sin kny' = b(x - x')6(y - y') (B.28) rr n,n or -sink,nzsinkmz' = 6(-zx') (B.29) E sin k,ysin k,y' = 6(y -') (B.30) which can be shown independently. The particular solution is then j2p. e-jk,(z- z > z' A, =- jbk sin5k,,z sink,nysink,mz'sink,y' (B.31) m,n ~jk,(z-z') z < zt B.2 Field Behavior - Electric Current Now that we have the particular solution for the potential, our interest turns to the behavior of the fields, particularly near the source. The I field is determined by evaluating Equation (B.4) which yields Z2k1 e) z > z' H~ = -C j a', sin k,mz cos kny sin k,,,z' sin k,y/.... (B.32) m,n~~~~~~~~~ e -,~'-"

iSO Z LE. COSkmZ sin kny sill k,m sin k,nY' 3\ j2k,n f e m,n ka(:-:e') z < Hz= 0 B.34 The E field can be found from (B.12) with the Lorenz condition anyd is Sikm kzikysn xik_- jkiz-:) z > EXr = Eaw cos k, sin ky y sin kmz'' sin ky' (B.35) *( 2kn')(_-k)(z- zz')')> ( Thus,Ey = fin sin kmt cos kny sin km c' sill kny' ( B.3 ) m,n kab (z- ) z3), j2w1 f nc sin kor s in ak uh sin ke c Tsh te Ahus, waspevfiousl pntecut, t lunction ins noral ly plie to (Th p of an Zt 2. sink.,zxsinkk,yYsinkk,,z'sin k,Iy' i(z- z') m,n ab Ck, (z-z' ) 1, a b 2 I s 2 + k 2 eeJk. ( r - r') Z < -' _(.._z- z')6(yk- y')6(z- z') > Thus, we find that the particular solution contains not only the normal (TM) waveguide modes but also a singular term in the i component of the E field, that is, the addition of the term i(z -')6(y - y')6(z - Z') (B.38) This addition corresponds exactly to the singular term previously found by the vector wave function formulations discussed by many authors (see especially [78]). This term corresponds to the,i term associated with a dyadic representation. As was previously pointed out, the function is normally applied to the tproblem of an F( R) = /jE,ca. |II) )dVA (-39)

lSI where F is a field or potential and GF is a dvadic Green's fuiiction of tile appropriate tVpp In thle case where F is to be a potential, the singular term does not appear directly but should appear when the E is found by carefully performing tile second derivatives indicated by the'VV. A operation. Similarly, if F is to be the E field, the correctly constructed dyadic Green's function will be of tile electric field type and, in this case, will have the form E = -~..6(z- z')6(y- y')6(- - z') GCE = -Z.)+ Go (B.40) jwG where Go is a modal expression. Finally, we note that tile solution found is the complete solution for J = z6( R - R') in rectangular waveguide wilicih can be seen to excite only the TM modes. The TE modes can be found by assuming E = V x F whiich are excited by currents in the other two directionis. The point also illustrates the benefit of the vector wave function approach with dyadic analysis. Bothl the vector potential method and field expansion methods employing auxiliary potentials, when performed on a vector level, require each component of current to be considered separately. In contrast, the dyadic analysis produces the entire Green's function at once. Since all of these methods contain, the same pitfalls when the derivatives at discontinuities are not carefully performed, this is the only fundamental difference. Of course, depending on the structures involved, when other beneficial techniques such as the use of impedance boundary conditions and scattering superposition are not employed, the difficulty of the various derivations may increase dramatically. B.3 Potential Theory Solutions for Magnetic Currents The necessity of the singular term is particularly evident in the literature pertaining to the analysis of apertures in rectangular waveguides and cavities. In these cases, the problem has a'dual' nature in that we usually represent the apertures with magnetic currents I l,

182 dual problem can then be handled in exactly the same manner as follows. We take as the dual form of Maxwell's equations: V x E = -jw -(z - z')(y -')6(z- ) (B.41) Vx H = jwcE (B.42) where we have introduced the magnetic current source Ki = i( - z')i(y - y')(z- z') (B.43) Since V E = 0 for the dual case, we set E = -V7x F (B.44) and search for solutions to V2 + k2r= EK (B.45) Because K has only a z component we set F= Fzi (B.46) reducing Equation (B.45) to V2F. + k2Fz = E(z - x')6(y - y')6(z- z') (B.47) When F, has been found, the fields will be given by E = -V x (F,i) l OF, laF, -1Fs. 1F —Y (B.48) cdy at VV.(Fzi) H = j,,,Fi- WVV( i- # [ar + ya2 F.1 a[1 82 s] (B 49 ) jwpc [axz 8z jwp Oz2

183 Following a procedure identical to thle previous case, thle particular solutionl is found to be FZ = - - Cos k,k cos k, y cos k,' cos ky' cos (:; B.50 n,ejnk,(z_- < where cm and,n are the Neumann numbers defined as 2 i,f0 et =;o=(B. 51 1 i= 0 Note particularly in this case that for m = O and n = 0. the potential function is non-zero, unlike the previous case, and is the term referred to as the (0, 0)'mode' in [86]. However, as can be seen from the field expansion below, this mode does not make the key contribution at the source, but rather, the additional contribution comes from the singular term. B.4 Field Behavior - Magnetic Current The fields produced by the infinitesimal magnetic current source are the following: The E field is determined by evaluating Equation (B.48) which yields Er = - 2k,ab cos kmz sin kny cos k,z' cos kn,y' J > (B.52) Ez = O (B.54) The H field can be found from Equation (B.49) and is f Cnkm_e ejk'(z-z') > z' H = E j1 sin km,z cos kny cos kmz' cos kn (B.55) Hy= M COs kkmTZ sin kn cos kz' cosknsy' (B.56) ~~m,n~ ~' fe.( z-e () Z < z'.= Z~ ( k-j absm cos kmX cos km y cos k,y z cos k,y- t ) _ E m >n w.s kmz cos kny cos kmZI co( kny'diz < z' m$:~, ~,~ 2~pabcossk,~y coskyok' cosky'( z IWILE~ ~ ~ ~ ~~~~~~Js(-' Z G Z

184 k, 2ab COSkm z cos k Y cos k; z' cos k, y en Y' _6( - )6(y - I')(y - )- - ~B.5T jwLB Again note that these solutions are complete for the z component of current which excites only the TE modes in the waveguide. The remaining modes may here be found by setting H = i V x A with A = A,i in a manner similar to the electric current case. For the problem of a slotted waveguide, the singular component which appears in the i component of the H field has the form - b(z- z')b(y - v')6(z - Z') H = -z (B.58) An integral of the type H( R) = JJJCH(R, R) K(R)dV' (B.59) is typically applied to find the H field in the guide for which CH( R, R') has the form GH = z(z -')b(y - Y')i(z - z') H -iz+ +Go (B.60) It is obvious that the singular term will significantly alter the numerical value of the'selfcell' and neighboring cells (with overlapping basis functions) in a Method of Moments solution to a problem of this type and therefore causes serious consequences when overlooked. B.5 Partitioning It has been- observed that the singular term appears above in the z component of the field as a result of partitioning the field about z = z' when we applied the radiation condition. In fact, the occurrence of this term also depends on the choice of modes used to expand the field. For example, for the multilayered shielded microstrip problem discusse.d

185 in Chapter IV, it is far more convenient to express the Green's function inl terms of LSE and LSM modes and partition the problem parallel to the planes of the dielectric interfaces. Although the radiation condition was not iniposed on that solution, the singular term appeared for the component of current normal to the interface for the normal field. Imposing the radiation condition in that case, amounts to choosing a contour for the Fourier integral which does not produce additional singularities. It can thus be seen that tlhe singularity does not depend soley on the imposition of the radiation condition, but moreso on tile combination of partitions and the manner in which the fields are expressed. Tlerefore, we can avoid the appearance of the singularity in certain components if necessaryv hby careful formulation, now that we understand the origin of the singularity. These observations, however, raise a possibility which has not been addressed. Thirotughout this work we have emplhasized the simplicity in the application of boundary conditions which results from the use of the LSE and LSM modes to expand the fields iM variolis structures. This practice also avoids the singular term in the Green's function, since it appears in a component which is not needed for the planar circuits treated here. Hiowever, it is not clear that this is necessarily the most desirable form from the point of view of convergence for the modal sums. It may be that the explicit extraction of the singularity may improve the convergence of the remaining modal sums for that direction. If this is the case, it may be more desirable to retain the singularity in the direction of the current being modelled, even though enforcing the boundary conditions becomes significantly more difficult analytically. This question should be considered for future work.

1S8 APPENDIX C On the Use of Vector Potentials with the Method of Scattering Superposition For a magnetic source Maxwell's equations take the form Vx E = -j H - K (C.1) V x H = jc E (C.2) ere the constitutive relations D = (E and B = pH have been assumed. Since from uation (C.2) V. E = 0, E can be expressed in terms of a vector potential function F as E= -Vx F (C.3) Using Equation (C.3) in Equation (C.2) and the usual Lorenz condition [82J, F must satisfy V2F + k22F = cK (C.4) and consequently, = jwF +!7VV.F (C.5) For an ~ directected infinitesimal magnetic current in a homogeneously filled infinite space, K = i6(R- R') (C.G) so that Equation C.4 becomes V2 F + k2 r = 6(z - z')6(y - y')6(z - z') C

187 Tile particular solution in this case is knowni to be'P) _ ( F = -t kR ( e where R = /(z - z')2 + (y - y')2 + (z - z')2. Sommerfeld has derived an integral form for this solution [69, Ch. 6] wliich is e-JkR A' AdA f0= AdA > (: JR = L Jo(Ap)e-I I = Jo(A)p)eU- dA ( 9) R u u where u = VA2 - k and k2 = w2pE. We now divide the space by introducing an infinite, perfectly conductiiig ground plane below the source and a dielectric interface above. This introduces new boundary conditions which must be satisfied. To accomplish this, we are free to add solutions of the homogeneous form of Equation (C.5) (i.e.,'source free'). These are taken to be in the forms: F+ = — 4 |A+(A)Jo(Ap)e-"z z < h (C.10) F;- -, [ A()JO(Ap)euz z < h (C.11) -: =-o |f A-(A)Jo(p)e-uozd- z > h (C.12) Uo in which A and AO are to be determined so that the new boundary conditions are satisfied. These terms can be interpreted as producing plane waves scattered from the ground plane and interface with components travelling in the ~i directions within the slab. The terms with superscript'o' corresponding to fields above the interface. Sommerfeld has shown that we must also include scattered potential functions having e components in order to satisfy the boundary conditions'. These are taken as A2dA FZ= -- - cos 4 A+(A)JI(Ap)e-' (C.13) 4 o iu F- = 4- cos4 | A'(,)Jl(,p)euzAtdA (C.14) 4 cos 2d Fz~= -4 cos A-,(A)Jt(Ap)e -'AdA (C.15) 4~r o (C. 14'This choice is not unique as noted in [(25]

s8S where uo = v/-, k2 =;2o(ok and o is tile angle -p makes with the z axis. It shlould be noted that the functioiial forms hlave been chosen in anticipation of derivative operators which will be applied wlien the fields are evaluated at the boundaries. The solution in the slab then is F = ( + I(F+ + F-)+ -(F+ + F-') (C.16) while in the infinite space above F = ~F2 + F~ (C.17) In view of the expansions of Equations (C.3) and (C..5),. ~OFZ.Y (%F, aF,) iFZ (C.18) X E)y a dz Oz e ay H = k) [5 (F++. Y V F+ kFz + rV F)] (C.19) We apply the following boundary conditionis at the interfaces: zx E=0 at z=0 (C.20) x E+=x E- at z= h (C.21) ix Hf =2ix H- at z=h (C.22) Here the ~ superscripts imply evaluation of the field on either side of the interface (not to be confused withi the ~ waves used earlier). No additional boundary condition is needed at the source plane, since the particular solution already satisfies the source conditions and the scattered potentials are solutions to the homogeneous equation. This leads to the following system of equations: A+ As = O (C.23) A - AZ = e" (c.24)

189 i[Az e + Az euhj = 40e-oh U 110 A u h A- uh- AO,usuh - e-.(- C 7'.2) + Ae- uh c -uoh - -u(h-)C. -(A e-u - A(u AO Ue - ) (".2 C.27) 1IA.o o( -uh h (hh 1') -[Ae-u + A -eu] + [Ate- - A-eu = c Aseuoh + A.e-Uoh -es( which when solved produce cruo sinhU(h - z')+ + coshu(h - zU)] 9 A+ - rcu z sinh[u(h - z')] + u coshu(h - z')j - z (C30) I fi(r, u, u0, h) = U(Cr - 1)cosh(uz') A,+ -A-(C..I whh fl(r, u, uhol, h)f2(u, u,h)C AO = elsh2ErUO cosh(uz') (C.32) Z fl ('r U, uo, h) A0 = Uh 2u(e,. - 1)sinh(uh) cosh(uz') (C 33) fl (erfl(, uo, h)f2(u, uo, h) f (c+ u,u0h) = r ucosCSh(uh)+ usinh(uh) (C.34) f2(u,u=,h) = ucosh(uh)+ u0sinh(uh) (C.35) We then find that cosh(uz') [Cruo sinh(u(h - z)] + u coshlu(h - z))] 2wr U f1 (C, o, Joh) Zz'' (C.36) F.0 f" AJ,(Ap) cosh(uz') - uhdA z > h (Ch37) F2 —1 f1(cuh e"'dflu, uo, hz F2 -A - I ) =o _eJ1 2uo(Ap) sinh(Uh) cosh(uz') 2w COS, Jo f dA z<h (C.38) (E7 - fl). Uo, h =2Ji(.p)sinh(uh) cosh(u') (C.34) F2- 2w cs(U], fUoU,h)f=(u h) d zin(h) (C.3))

190 These expressions are identical to the results in [37] fron which the fields can be found ti. applying Equations (C.3) and (C.5). The fields correspoiid to the ii, z i. yz- componcnrt of the dyadic Green's functions. For - and directed currents, the process must be repeated (although the y terms can be obtained by a coordinate rotation due to svmmetry').

BIBLIOGRAPHY 191

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APPENDIX B Norman L. VandenBerg and Linda P.B. Katehi Broadband Vertical Interconnects Using Slot-Coupled Shielded Microstrip Lines

IEEE TRANSACTIONS ON \MICRO,XA,E THEORM AND TECHNIQOES %OL,0) xO I JA\L ARE:, Broadband Vertical Interconnects Using Slot-Coupled Shielded Microstrip Lines Norman L. VandenBerg l.t'heer. IEEE. anti Linda P. B. Katehi. Steri,r.ht, /,t'r,-. IE.'E.Abstract —A full-saie space-domain integral equation anal-.sis for microstnp-to-.,lotline has been reported in [,0 Ik ith,sis of aperture coupled shielded microstrip lines is presented applications to open structures..A quasi-,tatic anal i-, ha, based on the Equi,alence Principle. The formulation includes been pro,.ided in [10], hov,e,er. thi. nia, no~t he Nuttlthe capability to model multi-lavered substrates through the derivation of the associated dvadic Green's functions,which cient particularl for higher frequenci-' here end errepresent the layers through impedance boundary conditions. fects and higher order mode coupling become more,sitThe method of moments is used to solve for the line currents nificant. A more recent paper [1 1] presents a transmlisl,on and slot voltage with even and odd mode excitations w-hich are line analvsis with excellent results. however. the method then interpreted through transmission line analysis to deter- does not account for radiation if the structure n. n'... s does not account for radiation if the structure 1> Open. inmine the two-port scattering parameters. A parametric study together with experimental data is presented which demon- teractions with the shielding if it is closed as in our casc. strates the behavior of the coupler and the accuracy of the tech- or other electromagnetic effects which become more senique. vere as frequency is increased. Hybrid methods wAhich combine two-dimensional full-wave analvsis w ith transmission line theory. as in [121]. should certainli extend I. INTRODUCTlON' the validity of such models. however. may still not acRANSITIONS from microstrip to slotline have long count for all discontinuity effects. especially since it is 1 been recognized as important circuit elements. Two usually desirable to minimize the overall size which!ill such transitions can be combined to form interconnects tend to allow transitions to interact. Our approach uses a between lines and by using lines on opposite sides of the three-dimensional full-wave space-domain integral equaslot plane. a vertical transition is made. The basic struc- tion method employing the method of moments i ith Galture. in a variety of forms. has a wide range of applica- erkin's procedure to account for all possible interactions. tions to both broadband and narrowband connections and can be used as a building block for interconnects [1]. II. INTEGRAL EQUATION FORMULATION AND NET\VORK phase shifters and inverters [2]. directional couplers [3]. ANALYSIS filters [4]. and many other microwave components [5]. The basic structure of the coupler to be discussed is as The advent of monolithic techniques for microwave and shown in Fig. 1. Variations on this geometry include cases millimeter-wave circuits has amplified the need for ac- with microstrp lines on the same side of the slot; multicurate analysis techniques to account for the effects of layered substrates/superstrates: reverse couplers where the shielding structures as well as the interaction between cir- lines exit on the same wall; and with additional parallel cuit elements which may be closely spaced. These effects slots and lines, among others, but can all be analyzed usmay not be accounted for by some methods such as simple ing the same approach. transmission line analysis. especially as frequency in- The analysis proceeds as follows: the slot is replaced creases. Additionally. high-frequency interconnect ap- on both sides by an equivalent magnetic current backed proaches are required which must be accurately modeled by a perfectly conducting wall, representing the tangential for design and also must fit well in a monolithic fabrica- electric field in the slot (all walls will be assumed to be tion scheme. In this regard. the presented approach has perfect conductors). The problem is thereby separated into significant appeal over via holes since it implements a independent regions, coupled independent regions, coupled together by magnetic curvertical interconnect while requiring only planar elements rents as shown in Fig. 2. This figure illustrates the treatwhich can be accurately modeled. ment of finite slot thickness by introducing an intern ening Numerous investigators have presented approximate cavity. Eliminating the cavity and replacing KL and Kt by analytical techniques to characterize these structures with K reduces the slot to the infinitesimally thin case which applications to circuit elements [4]-[8]. A full-wave anal- will be assumed here. Using the same current on either Manuscnpt received March 1. 1991. revised July 10. 1991. This work side enforces the continuity of the electric field in the slot. was supported by Texas Instruments under contract UM/EECS:390442. The fields in the cavities can now be written in term', oft the National Science Foundation under contract ECS:865795 I. and by the integrals as follows: NASA Center for Space Terahertz Technology. The authors are with the Radiation Laboratory. the University of Mich- -. igan. 1301 Beal Avenue. Ann Arbor. MI 48109-2122. E= -jo,,i G, *J dS - iG A /S 1 IEEE Log Number 9103898.ee 0018-9480/92S03.00 1992 IEEE

IEEE TRANSACTIONS ON %I1CROWA-,E THEOR' AND TECH'IQLES',)L - _e K J G.K KdS' -j, J, dS = )','c.-:\x".:x%"..xt,nlot,np, e' ~' "?/.:.... ~//; ///,/ /, where 0* implies that the tield is nonzero at lap.n t,locations. Equations (3) and (5) are e aluatcd,,n th i,,\' and upper stnps respectivelv. and (4) on the,1,lt TKhe r',scripts L' and L indicate w'hether the ourcce;, in trni rp.e or lower ca% it-. The longitudinal current components are no'k -\panfded in terms of piecewise sinusoidal functions,, xith a \la\wellian transverse distribution satisf\ in, the edoe conditions for narrow strips and slots. In general. the currents are written in terms of both longitudinal and trans,,erse components however, in this case. it aill be assumed that both the slot and the strips are narrow enouah so that the roDedance longitudinal components of current dominate their beha',Bo,;Inc. ior and the transverse components can be neglected. The Smp integral equations can now be written as a generalized z=d' matrix equation in conventional method of moments fashCavity ion. Using Galerkin's procedure. the matril is nearly E c.... KL svmmetric. the exception being the negative si_ns in the. -K off-diagonal quadrants representing the coupling terms Eca,.L K between the slot and microstrip lines: C -'V.L Lower z =d- ZLL YLK - KL \ Impded ~ ZKL = _YK YLL + YUU ZKL = TYUK'=d SCvImeacLK +- KL 0 oUn YUK = -Z C ZL'L' Fig. 2. Electnric and equivalent magnetic currents for microstrip-to-microstnp coupler with a "thick" slot.'L ~* VK 0= ~ (6) H = J GJ J dS' -jwe GmK K dS' (2) L/ U, LO* where T indicates a transposed matrix and the "*" indiThe subscripts on the Green's functions indicate whether cates the exceptions a the gap generators used for excithe function is of the electric (e) or magnetic (m) field type tation of the lines. In addition to the relations indicated, for electric (J) or magnetic (K) currnts. for a symmetric structure only three of the submatrices By using the appropriate rectangular cavity Green's are unique and the matrix will then have'mirror symfunctions, the fields n the cavities satisfy the boundary metry with respect to the cross diagonal when properly conditions, zero tangental elect mac fields ning boundary condi- loaded. The structures discussed here will be assumed tions, zero tangential electric fields n the strips and con lossless and symmetric for simplicity, although the more tinuous tangential magnetic fields in the slot, allow us to general case can be handled with the same approach. general case can be handled with the same approach. write the integral equations which are The inverted matrix is then used with even and odd gap I.YpSg.' _ generator excitations at the line ends to find the currents — j J GeJ JL dS' - G,K K dS' = 0* (3) on the microstrip lines. From the even and odd currents. stnpL slot even and odd impedances are found by measuring d defined as the relative distance from a standing wave maxima to the location of the slot. The expression for the reG,,.J. JL dS,' flection coefficient referenced to the slot then reduces to stnpL r = — e 4d/ (7) _ j j SI [eLGK + eU,GmK] KdS' which produces an impedance according to slot +r Z=.(8) - J Gm. JL dS' = 0 (4) The even and odd impedances are then combined to form,tnpc the Z-parameters which, because of the assumed svm

,, ANDENBERG -ND KAITEHI BROADBAND V'ERTICAL INTERCONNECTS metry are given by the simple expressions: 1- Z(, 9) Z - Z10 Z, = = Z. 1 1,-: Finalkl. the exn ind ()dZ impedances ma. be combined to produce S-param-neter, throuh the transtormations: - Z7 - Z - 1 =. ( 1) Fli. l Impedance voundarBn c,5nil!tn rerndi.nt.nat,n'...r.;. Z I- ZZlI- Z-l - I SI'u.,mture S, -=Sl', "IZ Z So! = -z 2Z - 1o l = Sln ( 12) pling coefficients for each laver. The solution. are C\panded in terms of LSE and LSM modes to tfacilitae vwhich are used to characterize the couplinc behavior. finding the fields in the remainina lax ers since these modes are decoupled on the boundanes and can be WndiIII. DERIV,A-TION OF THE DEY.ADIC GREEN'S FLNCTIONS viduallv matched to identical modes in adiacent l\crs In order to solve the integral equations. the Dyadic This also produces a one-to-one uniqueness in the ImpedGreen's functions must be known. They are solutions to ance condition for each mode. The coupling coefficients the dyadic Helmholtz equations: are then readily found by matching the tangential held components at successive layers. V x V x G, - k-G,.J = 16(R - R') (13) The impedance boundary conditions require the fields to satisfy v x v x Gm, - k-G,j = x [I6(R - R')] (14) i E i V x Gm,j v x x GmK - k-GmK = 1(R R =jREr. ( I v x r; x GeK - k-GeK = V x Ib(R - R') (16) For simpler notation, e, will be the wave impedance associated with the LSE modes and,,, will be used for the where the pnrimed vectors throughout. in this case R'. in- LSM modes. ke and t, will denote normalization to the dicate a function of the source coordinates. As can be wave impedances in each layer and are defined as seen. these equations are highly symmetric and in fact, once we have solved one set. the solution to the second k:. 7ei, W, - tn set is almost trivial. Note however, that while the second =, (18 pair of equations can be produced by application of the Duality Principle, the solutions cannot because duality These impedance boundary conditions derive from the does not apply to the boundary conditions [13]. The equa- recognition that the layered structures are simply sequentions may be solved by a variety of techniques including tial sections of homogeneously filled rectangular wavevector potential methods or. as we have done, through a guide sections. We can then evaluate the impedance confield expansion method which represents the solutions in ditions using transmission line analysis and the wave terms of vector wave functions (VWF's). This latter ap- impedances for the various layers as illustrated in Fig. 3. proach has been developed to a straightforward procedure The impedances on a given layer upper boundaryv are employing dyadic analysis to produce all components of found by the transmission line equation the dvads in one exercise. For the sake of brevity, only the final results with be given here. A more detailed de- k., + j tan k:,_ l,,, I nvation is available in [14]. ei k-_ I i + jm, - tan k. =, - I, (19) The Green's functions can be generalized to represent any layered structure of the type illustrated in Fig. 3. The use of impedance boundary conditions allow for the treat- Ek (20) ment of multi-layered substrates and superstrates as may e(_,i1ke} LI + jI,m(_-,) tan k:,_,,),,_7j be encountered in monolithic or other microwave circuits. The procedure is to find the Green's function for the layer where the index i is iterated from the top wall (rT.o = rh,,o containing the strip or slot by applying transmission line = O) through successive lower layers to the layer of intheory to the other layers to obtain the impedance bound- terest (li is the thickness of the ith layer). A similar proary conditions (rt) corresponding to each mode. Subse- cess is used for the lower layers in which case the iteration quently, the fields in the other layers can be found from proceeds from the lowest layer upwards and the wave the homogeneous solutions to (13)-(16) and a pair of cou- impedance is negative.

IEEE TRANSACTIONS ON MICROWAVE THEOR'' AND TECH%IQLES.OL',',, \:' The final results can be written as 6 (,[i7mr: k,. k,. k.z - c)l kt,'l,,,[:. k,. k. k- 1, - _- 6.... k d k,. k.. k z-.- abk.k - c1, OSkc - ) os,,L -- j), 7,in k. - ii ([,,[,.' kT. k... k. - c)I "t,4[71t' kr. k,. k:-z' d) 1,,L[7, L k. k. k.-( - d)1;,[[L' k,. k,. k. (z' )l k-ic r 7t;;r - d c;[1)e r -d sin for - -' 21,7'et- l,cL ) COS ktC - d) -J(,ec,LteL - 1) sin k.(c - d}) J and 2j(2 - 3,,m,,) G,e = I bR - R')+ C 2, =o n=-o abk:(kf + k:) {(lee[MeU; kt. k,, kz ( - c)] Mt,;e[i,eL; k,. k. k.(z' - d)] k,L k,, k, k -d)] te[Ii,;;. k,.. k-;(' - c)] (,)eL -,eL ) cos k:(c - d) - j(,.cc,eL - 1 ) sin k: (c- d) moo[,mU; kr, k,. k:(z - C)] ootOJimL; kr, k.,, k.(z' - d)] ( oo krmL;, k,-, k- ( - d)] 0o(i[7mu: kr, kv. k.(.' - c)f ('mL' -,tmL) cos k.(c - d) - j(,tmU,tmL - 1) sin k.(c - d)) where c and d are the upper and lower coordinates of the source layer. The,,,,, terms is the Kronecker delta defined chosen to satisfy the boundary conditions on the' alls of as b,, = I for m = 0 or n = 0 and zero otherwise. As the cavity and are given by indicated, the top lines apply for Z >'and the bottom sin k,x sin k v sin k. lines forz < z' where primes throughout indicate func- (0029) tions of the source coordinates. The functions 31Z and 9t e. e cos k,x cos k,.y cos k:are defined by 9,[r/. a] = /M,.,[ta] /+ jM,o[a] (23) - sin k,,x sin k,y cos k. e, o kcos k1x cos k,y sin k.z 0t,,,[?7. ac] = tqN.[caC] + jN.u[aJ (25) with k1 = m'/a, kv = nh/b and k. = k - k,, - k,. The subscripts here and on the M and N functions indicate te,.[nt, la] = ilN,,e[a] - jN,o[aO]. (26) whether the trigonometric dependencies are even or odd -- - (cosine or sine, respectively). The vector wave functions M and N are defined by (cosine or sine, respectively). The solutions for the equivalent magnetic current K are - I virtually identical with the following notational replaceM = V x *i, = k x N (27) k ~xI~i, k N(27) ments: I _ M * N (31) N =I V x V x = Vx M (28) (3 k k G,j G.K (32) wherc k: = k; + k; + k., * is a scalar function and.i G,1 - G.K. (33) is a unit vector called the'"piloting vector" which determines the nature of the field expansion. In our case, Note that the M * N replacements are what make these choosing x, =', the normal to the layer interfaces, results substitutions different from those dictated by the Duality in a correspondence between the leading M and N VWF's Principle. It is these substitutions which compen,ate for appearing in G,j and the LSM and LSE modes, respec- the change in boundary conditions from the Dlrnchlet to tively. (In contrast, it should be noted that leading M and Neumann conditions when the Duality Pnnclplc. apN functions correspond to the LSE and LSM modes, re- plied. Expansion of these functions then yield all hc cornmspectively. when they appear in G,j). The forms of ~ are ponents of the various Green's functions.

A\DENBERG AND KATEHI BROADBAND ERTICAL INTERCONNECTS IV. NU!MERICAL AND EXPERIMENTAL RESULTS oo:: A coupler with the geometnr of Fig. I was analyvzed: _ usino the above techniques. The parameters which can be vaned in this design are numerous. consequently. oniv a / 60 / rte% aanations \,ill be presented here. In all cases. al- - t though not required In oeneral..,mmetric geometrx is maintained to.mptilt' the e'.en and odd mode anal\sis. _ - as discu..ssed abov\e. A..l,k. in all cases the cross-.section > tor the ca it, ().25. (0.25 in.: the substrate is 0.025 5 in. thick with e. = 1) 6: and the,lot and line widths are, — 0.025 in. The cavit\ len,th is varied for the numerical results and fixed at 2.0 in. for the measurements. This,,,:,o,- dimension does not affect the results since for all frequen- Line Stub Length ir.,hes cies considered here. the cavity is below the cutoff freFig. 4. Effect or the line Ntub leneth /, on S, nd. S,-n;xuc, quency of the higher order microstrip modes and the ref- L = O 2 Inches and t = ()GHz erence plane was taken to be at the slot. To illustrate the behavior of the coupler. we first examine the influence of various parameters at fixed operating frequencies. The effect of the line stub length (1) is shown in Fig. 4. It can be seen that the stub is initially too long for an ideal match at this frequency. However, open' as the stub is progressively shortened. a certain length / jx ] "matches" the two port coupler and with further shortening the match gets progressively worse. We can interpret this effect by examining the equivalent circuit shown in Fig. 5. Variation of the stub length has the effect of open changing the position of the current maxima (virtual jx end shorts) and minima (virtual opens) on the lines relative to the slot. thus varying the degree of coupling through the Fig. 5. Equivalent circuit for a 2-port coupier slot represented by the coupling transformers. Consequently, the peak coupling occurs when line stub length 0.0 places a current maximum below the slot or lengths in odd multiples of - X/4. The opposite effect occurs when the 3 line stub is approximately in multiples of X/2 in length -6.0 so that there is a virtual open circuit beneath the slot, in which case there would be very little coupling between' the line and slot..'12.0 A similar effect is observed for variations in slot length / I (L,) as illustrated in Fig. 6. Again using the transmission 5 -1.0 - I. s line analogy. one can interpret this effect by transforming -18.0. the impedances at the ends of the slot to the center. These. end impedances are nearly short circuits. the difference -2.00.000 0.025 0.050 0075 0.100 0.125 0.150 0.175 0.200 0.22 )'; being due to fringing fields which extend beyond the ends of the slot line. fully accounted for by the full-wave anal~vsis. At the matching length. the resulting transformed Fig. 6. Effect of the slot length on S,, and S,, magnitudes i. = = 0.049 inches and f = 18.0 GHz). reactances at the center cancel the reactance associated with the junction. thereby matching the two ports. As the slot becomes very short. the field in the slot is effectively are varied to form a parametric database. (The most sig"short circuited," thus coupling is reduced. S-, then tends nificant computation time is in the generation of the mato zero while S,, approaches unity (since the structure is trix. This takes approximately 30-45 min per frequency closed and assumed lossless). All of these effects would on an — 25 MIPS machine such as the IBM RS6000, 320. be expected to repeat as the slot length increases in mul- One matrix is sufficient for all variations of slot and line tiples of X. however, for the case studied here, the max- stub lengths). The database is then scanned to assemble imum slot length is limited by the dimensions of the frequency response plots as a function of the geometric shielding package which have been chosen to allow only parameters. To verify the results, we have deslgned and the dominant microstrip mode to propagate. constructed the fixture shown in Fig. 7. Thi,,t\ur. JWTo generate a frequency response, the programs are run lows sample substrates with various line and,l.r.::,:nat each frequency of interest and the slot and line lengths sions to be installed in various combinations t,i,,jtll' —

IEEE TRANSACTIONS ON'11CRO.Ax E THEORr' AND TECH\IQL ES "'L-' - 0 -,^__ __ -YT- *; _ _ F,:';u:r.' GHFig 8 Comparison ot theonr,nd e~perimcent o,r.n.:xre,;ne.tuc t = 0. I = 0 11 in.. L, = 0)250 In. The crror bar:ndi,:i~ the.n,u un,,r Fig. Photograph of coupler tixture as,embl. the high frequency corner b? a -0 005 inch chin_e In stub en,_n quencv response measurements. A number of circuit tween the upper and lower lines introduces an asvmmetr\ boards were made in two sets: One set of boards was dou- which has not been included in the current numerica ble-sided with a microstrip line etched to certain lengths model and contradicts some of the assumptions stated fo relative to a slot etched in the ground plane on the oppo- the network analysis. A more general model is planned tc site side. The second set was one-sided boards with mi- include this capability as well. crostrip lines of corresponding lengths designed to be held The ripple in all the measurements can be shown tc against the boards of the first set by the fixture. result from mismatch at the microstrip launchers. The Measurement of one of the assemblies is shown in Fig. measurements are particularly sensitive to this connectior 8 in comparison to corresponding numerical results. The because of the high dielectric constant of the substrate position of the high frequency corner of the response was They occurred in varying degrees throughout our mea found to be very sensitive to the length of the line stub. surements and are also influenced by small air gaps be As discussed above. this corner is controlled by the length tween the connector assembly. fixture and substrates. Thi at which the stub is approximately X/2. Since the effec- ripple could probably be removed by more sophisticate( tive dielectric constant for the microstrip is approximately deembedding techniques however this requires additiona ceff = 7.8 at 17.0 GHz. a null is predicted in the response fixtures. Nevertheless, the ripple shown in the results pre in that neighborhood so there is good agreement with the sented here in not substantial and does not significantl! results shown. A theoretical curve for I = 0.115 inches interfere with the fundamental behavior of the devices is also shown which gives an indication of the sensitivity Also. the broadening of the low frequency response i to the stub length. The error bar on the high end indicates typical in the measurements. We were not able to identif the sensitivity of the high frequency corner to a +5 mil a direct cause for this effect, however, we suspect that error in line stub length which is well within the expected may also be related to the fixture/connector interface sine tolerance errors for positioning the stubs relative to the we have not de-embedded these transitions. We also pos slot, so we conclude that the results are in excellent agree- tulated that some of the anomalies might be caused by th ment. In fact, we were able to move the upper board side-wall grooves in the fixture which hold the double slightly toward the slot to expand the stub length some- sided board in place. This possibility was eliminated how what which did shift the high corner to a lower frequency ever by installing movable side-wall shorts which are vis as expected. However, this also created problems with the ible in Fig. 7. match at the Eisenhart microstrip launchers so these re- The remaining discrepancy is perhaps a slight addi suits are not shown. tional loss found in some of the measured results. T The "sidelobe" which can be seen at the high fre- deembed the losses for the structure, a through line wa quency end, is also attributed to tolerance errors for the measured and the remaining measurements were post-prc line stub lengths. A difference in lengths would produce cessed to compensate for conductor and dielectric losse multiple -nulls in the response at the high end which would on the microstrip lines. This process however does n( be expected to have a sidelobe inbetween. Because of the correct for losses associated with the slot including bot high sensitivity to line length, owing in part to the high conductor and dielectric losses and additional losses o dielectric constant, the amplitude and span of the sidelobe the cavity walls, plus losses due to the added line length! is a strong function of the relative line stub lengths, which The remaining differences are thus attributed to these fac can also be observed when the boards are slightly shifted tors together with measurement errors and are judged a as described above. The sidelobes does not appear in the be within acceptable limits. theoretical result since a difference in stub lengths be- Measurements on a different line stub length are show

-"\DE,BEFRG q\D K.TEHI BROADBAND VERTICAL INTERCONNECTS _ I, - ~.~ - ~.,. -- -;. _ Frrtuen,:,, tGHz) H,-. ii. Fig q Ciompir:,,,n't experiment jnd theor, torS, magnitude, ith s = Fg. 11. Comparison,noSlinC the e'cct m,, Ine,ep.ir.ii[;.\,'. (). / = 1) 080 n. L ().250 In. llustratin the control ot the high tre- In.. L. = 0 50 in jnd \ = I) "'n quenc c`orner kith the line stub length b, comparison with Fig 8. 0S U hi - - /S m 1 _ 1 - I' -12Q. -18.0 -18.0r i -21'...0 0.0 30 60 9.0 12.0 15.0 18.0 00 30 6 0 90 10.- Frequency (GHz) Frequency (GHz) Fig 10. Expenmental and theory for S, magnitudes with s = 0. 1 = 0.110 Fig. 12. The effect of line separation vith shorter line stub lenctn. in. L, = 0. 153 in. showing the effect of shortening the slot length. 0.080 in.. L, = 0.250 in. and s = 0. 125 In. in Fig. 9 again showing good agreement with the theo- V. CONCLUSION retical results. In this case, the shortening of the stub A set of integral equations for aperture coupled shielded length has moved the high frequency corner out of the microstrip lines has been introduced based on Green's range of the measurements. What is interesting to note function integrals and the Equivalence Principle. The asabout this case is that the low frequency corner of the sociated dyadic Green's functions in the form of uaveresponse is quite insensitive to this change in stub length. guide LSE and LSM modes have been derived uhich alAlso of particular interest is the wide bandwidth of this low for a full-wave analysis, accounting for all transition. electromagnetic interactions of the microstnp- slot couFig. 10 illustrates the effect of shortening the slot pier. including the capability for multi-lavered substrates length. One consequence is reduced coupling in the pass- and superstrates. By expanding the unknown line currents band which was also demonstrated in Fig. 6. We also see and slot voltage in terms of subsectional basis functions in this result. some movement of the high frequency null and applying the method of moments together with even due to a shortening of the line stub length. and odd mode transmission line analysis. the two-port The final plots. Figs. 11 and 12, show the influence of scattering coefficients can be determined to characterize the line separation parameter (s) on the frequency re- the coupling behavior. sponse. The figures correspond to Figs. 8 and 9, respec- The frequency response plots shown demonstrate the tively, which have s = 0. showing in general, a narrow- utility of the structure as an interconnect. With proper seing of the frequency passband as s is increased. This is to lection of the geometric parameters such as line and slot be expected since we have now introduced an additional widths, lengths, line separation. substrate heights and malength parameter which can influence the response through terials, the frequency response can be tailored to nt\ e the its relationship to wavelength. Here again, the numerical required center frequency. bandwidth. shape. e lt A has model is judged to have correctly predicted the coupler been shown, very wide bandwidths can be atChIC \ <'i \' hich behavior after the experimental artifacts are considered. makes the structure very versatile. Avoidance o. I,h(ic

88 ~~~~~~~~IEEE TRAN\SACTIO\S ONs`MICROVAA' E THEMR AND TFCHNQ \l, F - V for RF transitions and their inherent limitations by the use 131 R E Colin. Field The,,r,t G,::... t.\.I..... pp2-3 of planar structures to form vertical interconnects. to-. L andenBer. Full- nl.-;- Lether with the ability of the model to accurateli predict nL and enuplerr. F Ph D dF i jcjtf,n T'..nT.. the coupler beha, ior as demonstrated b, experimental re- Ann Arbor. 1 Q1 sults. are especiallY important considerations for design of monolithic circuits. In addition. although the approach discussed here ues, ertain simplif. ing a-sumptions about the symmetr\ ot the structures. the technique can be readily adapted to the general case. REFERENCES Norman L. \ andenBer s....\ - \I,~ N I % i' r,ic B 3 L E \,;, 111 R H Jansen. R G Arnold. and 1. G Eddl.,on. A comprehenslo -- Ph.D iegree trom; n, >Aknn.Arbor in I Q{0. iqi ~c',;'p.... CAD approach to the design ot' MMIC', up to %lI-v.a,e trequen- Ann Arbor n I. n - cie.s. IEEE Trans. Mtcromuave The-;rc Te(h ol 6. ln) pp From De,emrner': u ic -, -, ~~~~~~~~~~~~~~~~~~~~~~emplo,,ed b%, the.Alted-l3en,;.\; Gu-,:.A.~_'',-:" 208-'19. Feb. 1988. emploedb the.AIiBn Gu.. s 21 K. G. Gupta. R. Garg. and I. J. Bahl. Microstrip Linie~,ltd S'iou,tliti3 - Di'ision-\Nishj%%ka Opcrn, inni wkhere he \,%orked on e\percr_denza"'. r.~.:,, Norwood. MA' Artech House. 1979. here he rkdn cpr:n J. [31 T Tanaka. K. Tsunoda. and M. Aika\,a. Slot-coupled directional milimeter-Aa%e spread pct:ru -um xiJr- Sin, couplers bet,een double-sided substrate microstrip line, and their ap-, then he has \.orkeu t ine Raj.tin L -.i..n< - plications." IEEE Trans. Mticrowave Theory Tecit..'ol 36. no 12. the lni\ersit% *t \Iichian n rumr._,. rtun. pp. 1752-1757. Dec. 1988. pp. 1752-1757. Dec. 1988. ng of electromagnetic tields c,ith applications t,, circut, afnU Jantenna, 141 E. A. Manani and 1. P Anos. Slot-line filters and couplers.'' IEEE Dr VandenBerg is a member ot' Eta Kappa N'u. Tau Br.c P; inc:n IEEE AP. MTT and EMC societies. Trans. Microwave Theory Tech.. sol. MITT-18. no. 12. pp 1089- IEEE AP. MTT and EMC societies 1095. Dec. 1970. 151 M. Aikawa and H. Oga%,a. "Double-sided MIC's and their applications." IEEE Trans. Microwave Theory Tech.. -ol. 37. no. 2. pp 406-413. Feb. 1989 161 S. B Cohn. "Slot line on a dielectric substrate.'' IEEE Trans. Microwave Theorn Tech.. vol. MTT-17. no. 10. pp. 768-778. Oct. 1969. [7] E. A. Mariani. C. P. Heinzman. J. P. Agrios. and S. B. Cohn. "Slot line charactenstics." IEEE Trans. Microwave Thevrs Tech.. vol. MTT-17. no. 12. pp. 1091-1096. Dec. 1969. - _ Linda P. B. Katehi 5S'8I-%S' —SM'9 re 181 J. B. Knorr. "Slot-line transitions. IEEE Trants. Microwave Theorn _ ceived the B.S.E.E. degree trom tne Nationa Tech.. vol. MTT-_2. no. 5. pp. 548-554. May 1974. Technical University ot'f Athens. Greece. in 1-' 191 H. Yang and N. G. Alexopoulos. "A dynamic model for microstrip-. - and the M.S.E.E. and Ph D deoree,r(om th sololine transition and related structures." IEEE Trans M4icrowave. ~ University of California. Lon, Angeles,. in 1981 anTheory Tech.. vol. 36. no. 2. pp. 286-293. Feb. 1988. __1984 respectivelv. 1101 5. Yamamoto. T. Azakami. and K. Itakura. "Slit-coupled stnip trans- -~-In September 1984 she joined the facult\ ot th mission lines." IEEE Trans. Microwave Theorv Tech.. vol.- MTT- 14..Electrical Engineering and Computer Science De no. II1. pp. 542-553. Nov. 1966. partment of the University ot Michigan. Ann Ar 1111 B. Schuppert.'Microstrip/slotline transitions: Modeling and exper- -bar. Since then, she has been involved in the mod imental investigatin. "IEEE Trans. Microwave Theory% Tech.. vol. eling and computer-aided desigzn of' millimeter 36. no. 8. pp. 1272-1282. Aug. 1988. wave and near-millimeter-wave monolithic circuits and antennas. 11I21 H. Ogawa. T. Hirota. and M. Aikawa. "New MIC power dividers In 1984 Dr. Katehi received the W. P. King Award and in 1985 th using coupled microstinp-slot lines: Two-sided MIC power dividers." S. A. Schelkunoff Award from the Antennas and Propagation Society, I IE'EE Trans. Microwave 7heorv Tech.. vol. MTT-33. no. I11. pp. 1987 she received an NSF Presidential Young Investigator Avkard and a 1155-1164. Nov. 1985. URSI Young Scientist Fellowship. She is a member of Sigma Xi

APPENDIX C Norman L. VandenBerg and Linda P.B. Katehi Generalized Full-Wave Treatment of Shielded Multi-Layer Structures Using the Vector Wave Function Field Expansion Method

Generalized Full-Wave Treatment of Shielded Multi-layer Structures Using the Vector NVave Function Field Expansion Method N. L. VandenBerg P. B. Katehi The University of Michigan Ann Arbor, MI 48109-2122 1991 Abstract A full-wave, integral equation method is presented which generalizes the treatment of multi-layer substrates and superstrates for shielded circuits so that any combination and number of layers can be handled without reformulation. A vector wave function approach with dyadic analysis is used to expand the fields, producing all components of the dyadic Green's functions in one solution. The approach makes application to the analysis of shielded transmission lines very straightforward including the calculation of the fields in all layers. Application to the determination of fundamental properties of strip transmission lines are emphasized in the examples provided.

I. Inltroduction Successful design of microwave circuits depends in part on the availabiit!y cG accurate analSysis and design tools to characterize transmission line structures. The trend towards higher frequencies has increased the need for analytical techniqlues which account for all electromagnetic phenomena in the structures, including tlhe effects and characteristics of higher-order modes. Mlany full-wave analyses haVe been presented in the literature to address these needs, including numerous papers on discontinuities in conventional microstrip for open and closed structures and. more recently, on slot and coplanar waveguide lines. As systems become more complex, so too do circuit geometries, including the integration of components on multi-layer substrates and superstrates. The approach presented here illustrates a full-wave method, which generalizes the treatment of multi-layer substrates and superstrates for shielded circuits so that virtually any combination of layers can be handled without reformulation. Although the concepts can be easily extended to the three-dimensional case for the treatment of discontinuities, the emphasis here is on the determination of fundamental properties of transmission lines which can be reduced to a problem of two-dimensions. To accomplish the full-wave treatment, the dyadic Green's functions for a multilayer rectangular waveguide will be found. The functions will be generalized so that they can be applied to a structure with any number of layers. The approach makes application to the analysis of shielded strip transmission lines very straightforward including the calculation of fields in all layers. As will be shown, the

process also makes available thle capability to visualize the fields in the structures. provi(iing insight on their electrical behavior. II. Derivation of the Dyadic Green's Functions The dvadic Green's functions needed for this case are the solutions to the dvadic Helmholtz equations: V x V x GeJ - k2Gej = I6(R — R') (1) V x7 x 7 mJ-k2- mJ = V X [6i(R-R')] (2) x V x GmK - k2GmK = I6(R-R') (3) V x 7 X GeK -k2eK = V x I(R- R') (4) The subscripts on the Green's functions indicate whether the function is of the electric (e) or magnetic (m) field type for electric (J) or magnetic (K) currents. The magnetic current types are needed in the analysis of slot and coplanar waveguide problems, however, will not be covered in detail here since the derivation closely parallels the electric current problem. With the solutions to these equations, the fields are given by E= -jjJEae,J dV-'fJGeK K dV' (3) H = JmJ J dV'-j jJf| mK R dV' (6) The functions L, M and N are vector wave functions forming a complete set of solutions to the homogeneous forms of the above equations. They are defined in

thle ~cncral fortms L 1 -, NI = V x TXi =-VxN xS - 1 1 - N = - x Vx = -Vx. x 19) PC K w3here re is a scalar function chosen to satisfy the boundary conditions of the problem, ii is a unit vector called the'piloting vector' and Pc is the separation constant 2 =* k2 + k + Ik2. Following the Ohm-Ravliegh method as described by Tai [1], we find the particular solutions by expanding the right hand sides of (14) in terms of the eigenfunctions L, MI and 4N with unknown vector coefficients; derive the value of the coefficients using the orthogonality properties of the VWFs; expand the dyadic in terms of the same functions with unknown scalar coefficients; and find these coefficients by performing the derivative operations and enforcing the equations. From Equations (7-9) it can be seen that only L can have non-zero divergence and since the right hand side of (2) has no divergence, the L function is not needed for its solution. It is therefore somewhat simpler to find the solution of Equation (2) first. Through Maxwell's equations it can be shown that GmJ and GeJ are related by V x GeJ = Gmj (10) V x mJ = I6(R - R) + k2eJ (11) so that GeJ can be found from (11) once GmJ is known. This is the essence of the'method of Gm' in [3].

Daladic Green's Function for Parallel Plate Waveguide Thie soluttion for the multilayered waveguide problem is built upon the dyadic Green's function solution of a parallel plate waveguide using the method of scattering superposition. The parallel plate waveguide is formed by infinite. perfectly conducting walls at x = 0 and z = a, filled between with a uniform dielectric material represented by the wavenumber k;,. where i will ultimately represent the Zh layer of a rectangular waveguide containing the source. The method of solution for this problem has been given by Tai [2] and in our notation GmJ can be written as 0 ~~ j(2-n )k 00 m=4rak,O(k2 k2 *[Mo(m. ky, ~kz)NO(m, -kY, Fk) + Ne(m, t4, kz)M(m,-ku, ikz)] 6> Z< (12) where k = /k2 _ k2 - k-. As indicated, the top sign applies to z > z' and the bottom sign to z < z'. The vector wave functions are defined by 0Io(k,) = V x o(k=,)z = V x [sin(-R)e-j(k,+ *)z] (13) a 1 1 nt nz ),-T (kyy+kz) ).Ne,(kx) = -V x V x T,,(k)z = -V x V x [cos(m )e-i+ z] (14) Kt K a The wavenumbers k, and k, are two continuous eigenvalues and kr = mir/a where m is an integer including m = 0 for Ne. Note that the'e' and'o' subscripts indicate the cosine and sine trigonometric dependence, respectively. GeJ can be derived by performing the operations indicated by Equation (11). This can be done almost by inspection using the relations between M and N from

Equations (S) and (9) except for the discontinuitS at which occurs at z ='..s in`:3] it can be shown that the relations given by Equations (S - 9) apply but an additional term is needed to account for the discontinuity at the source as in 00'- k~ j(2 - 6m)kl v x C, m = - _ dkz E J;(-5 I)k m = -J 4 z Takz (k2 + k2).[.Ne(m, ky, +k)-k).l'(m, -ky,:Fkz) +.o(m, kfl, ~k)N'o(m, -k,:tk)' + It6( R -') z z (1- where the transverse idemfactor, It = zi + yy, appears in this case as a result of the combination of the choice of the piloting vector z and the partitioning of the z dependence. The singular terms come from the derivative of terms of the form jk(z-z') z > z' f()- < (16) which is &f(,z) Zj e-jk*(Z-') z > Z' -_ _- jkz + 26(z-z') (17) ejik(z-z') z < z' This term has sometimes been overlooked with potential function methods as well. Thus, Equation (11) leads to 1 I\f0L(2 - 6in) GeJ = -'Z6( R- )- dk r + oo = —— O 4ra..(r y [,M,(m, ky, ~k,))M'(m, -ky,:Fk) + No(m, ky, I~k)NO(m, -k, tkz)] z' (1S) The solutions for the magnetic current are found with the same procedure. The results are identical to the above with the following replacements: GeJ i GmK, GmJ T G eK, and M: N.

Dvadic Green's Function for the Rectangular W\aveguide W\e noNw construct the dvadic Green's function for the source laver of a multilavered rectangular waveguide (see Figure 1) by using the parallel plate Green s function solutions above and the method of scattering superposition. Let Gmj = Gmj + ~mJ (19) = (P) 2(S) where G m is the parallel plate solution and G(i is defined as =(S) j(2-m)ki GmJ,=-J_oo 4 kdk. E 4raz(k2 + k2) [.(m, k, k) + (mk,-k) A + Mo(m, k, k) f3+ + M(m, k,-k) -] (20) representing fields which are scattered from dielectric layer interfaces located at z = c above the source and z = d below. The unknown vector coefficients, A+, A-, B+ and B-, are found by applying upper (77u) and lower ('7L) impedance boundary conditions for the layer. This approach greatly simplifies the analysis of the total structure by allowing the fields to be found first for the source layer alone, after which, expressions for the fields in the remaining layers can be immediately written in terms of the homogeneous solutions by mode matching. The boundary conditions require the fields to satisfy I = E (21) For electric currents, this becomes m = - or _= jVr x(2 2) 6r I.' ImJ

Sirnilarii for the equivalent magnetic current Green's functions. the impedance )ounlldarlV conditions are X GeK 2 -GeK jr/ = je or = = 2:3 yGmK yv x GeK For simpler notation, r1/ will be the wave impedance associated with the LSE modes and r/m will be used for the LSM modes. Furthermore, 0, and 0, will denote normalization to the intrinsic modal impedance in the layer and are defined as 1etm= (24) (25) kz; where i is an index associated with the ih layer. WVe can then evaluate the impedance conditions using transmission line analysis and wave impedances for the various layers as illustrated in Figure 1. The impedances on a given layer's upper boundary are found by the transmission line equations 7i= k [ e(i_i1) + j tank(_)l)] (26),kz(i,) 1 + JO7e(i-,) tan k((,_),)I(il) =7mi =fjk(i-j [ r 7m(-1) + jtanki_)li_] () (i-)kTi 1 + jm(i+)-) tan k.(i+l)l(+l) kzi, [ e(,,+,)-j tankz(i+,)l(I+l)] (28) kz(,+l) 1- 7je(j+l) tan k,(i+l)l(i+l) 0~,= (29) (i+l)kzi 1 - t?7m(i+l) tan kz.(i+l)l (+l)

Figure 1: Impedance boundary condition representation of a multi-layered structure. where here the iteration proceeds from the lowest layer upwards. Applying the boundary conditions produces two pairs of equations for the unknown coefficients: (?leu- l)e-jkc A+ - (,eU + 1)ejk'c A = -(,- 1)e-jkc'Mi(m, -ky, k,) (30) (f/eL- l)e-i'ad A+ - (/eL + l)eJk'd A- = ( /MeL + l)ejk'daMe(m,-ky, -kz) (31) and (0mU - 1)e-jk8c B+ + (Ij,U + l)ejk'B- = 1(imU l)e_jk'N(m,-kr, kz) (32) (OmL - l)e-jid B+ + (OrmL + 1)eikd B- = -(7OmL + 1)ed' -k,-k)(3) These four equations decouple into two pairs due to a judicious choice of the piloting vector in the VWFs. Because the piloting vector was chosen as the unit vector normal to the layer interfaces, the M and N VWFs appearing in GmJ

coirespond to the LSMI and LSE modes of the waveguide, respectively. (In contrast. M and N correspond to the LSE and LSMI modes. respectively, when they appear in GeJ.) As is well known, the LSE and LSMI modes are decoupled on the interfaces. that is, the tangential components of an individual mode on the interface can be matched by a like mode in the adjacent layer, therefore the coefficients are decoupled. That there are only four equations is a result of the fact that the field in a given layer depends only on the field in the adjacent layers. The solutions to Equations (30 - 33) are easily found and when combined with Equations (12,19) and (20), and some algebraic manipulations lead to [o oo j(2 - 6m)ki Gmi dk, — oI = J dkog 2-rak (k2 + k2) (Mo[ijmu; m, k,(z - c)] "0[0mL,; m, k,(z' - d)] (7Imu - T,mL) cos k,(c - d) - j(imUt,mL - 1) sin k, (c - d) ~(.e[([; m, k,(z - c)] 1[,L; m, k(z' - d)] 1 fe[TeL; m, kz(z - d)] AM,[i,eu; m, k,(z' - c)] (ifcU - Ti,L) cos k,(c - d) - j(i,crUieL -1) sin k,(c - d) for z z' (34) where we have defined the Me and N%0 functions for the sake of compact notation: Mo[77; a] = 7rMlIoo[a] + jMoe[a] (35) Je[7/; =] = /77Mee[H] - jl,eo[[ ] (36) fo[7; oC] = -tNoo[Q] + jNoel[ r] (37)

\e[77a] 7= 7Nee[] - jNeo[ot] Z' Since a here is [,m. k,(z- < c.,d >)] and m is associated wvith k,. we recognize that the'e' and o subscripts in this case imply trigonometric functions of x and z. Also note that the primed functions use -k,. WVe can find GeJ as we did in the parallel plate case, by performing the derivatives indicated by Equation (11): GeJ = -zz&( R- R )+ dkby,- 7327rakX,(k2+ k2) m ~[,; mr, k.(Z - c)] M.'[~,L; m, k,(z' - d)].t,[ e,,; m, k,(z - d)] M'[,l,;,, k,(z' - c)] (ieU - t~eL) cos k,Z(c - b) - j( OeUeL-1) sin k(c - b) ko[cm[; m, k(7C mL; m,( - d)c 1 Xfo[,,mL; m, k,(z - d)] [imu,; m, k(z' - c)] (inmU -'7mL) cos k, (c - b) - j(imUtmL - 1) sin k, (c - b) for z<z' (39) Both GeJ and GmJ involve a spectral integral which can be reduced to an additional mode summation index by Cauchy's Theorem, once the impedance functions are specified. For two-dimensional problems, this is not necessary as will be shown. 10

III. Generalized Application to MIulti-layered Shielded Nlicrostrip.Altlhough the derivation of the dyadic Green's functions using the approacn above hlas particular advantages when all of the components are needed. the applications to be discussed in this section will be restricted to one current component to minimize the complexity of the presentation. The objective for this section is to show how the fields in the various levels can be calculated for a known form of current on a single microstrip line in a straightforward way. There is no restriction on the placement of the current or on the number of strips which can be used. however, a simple case here will better serve to outline the method. The procedure for the treatment of more complex multi-layer coupled strips can be found in [4] which also is an example of the use of potential functions with impedance boundary conditions to generalize multiple layers in a similar way. In order to calculate the characteristic impedance, the definition Z = P/I2 is used where P is the time average of power propagating along the guide. Thus, if we set I = 1, the characteristic impedance is simply Z = P which can be computed analytically by integration of the average Poynting vector on the cross section of the waveguide. For our purposes, let us assume a geometry such as shown in Figure 2 with a longitudinal current component on a single narrow strip of the form I IOt 26(z' - d) J(x', y'') = -- e('kgv 1d' - x01 < W/2 (40) irW i -4 (</)2 i.e. a Maxwellian transverse variation of current which satisfies the edge conditions. The k9 parameter represents the set of characteristic complex phase constants

z=d:i:'.:;,,.'.:::.-.;.:: *::...............!. l R,;-C'.'.~:;".:.:'':................!:::X 0 x =a Figure 2: A shielded stripline, uniform in the y direction with multi-layered substrate and superstrate. associated with shielded microstrip modes. They are the eigenvalues of Et = -jwy jJ h X Ce J dV' = x Z,. J (41) which enforces an appropriate boundary condition on the surface of the microstrip line [4]. A particular value for k2 must be found first, before the fields of a particular mode can be calculated. Equation (40) is used in Equation (41) with the yy component of GeJ as derived abov,e and Galerkin's procedure to enforce the boundary condition. On recognition of the sifting property and integral representation of the delta function: _ e ~-(k'- )Y'dy' = 2r6(k - ky) ( 12

we fin(i tlhat tlle weighted l component of the electric field can be written as (2- k-I' E - a _=o k((k~k) sin2 k'xoj(').k2 L[ L[7[ COScos k=(c-d)(+ jsink (c-d)] c - L -l, [,.) cos k.(c - d) - ) sin k.(c - d) k.2k,2 -,)Cos k,(c - d) - j(', - )sin kkz(c (- d3 + z g I] (43) k2 [(mU 7mL COS k (cd) - d) - 1(mU?1mL - 1) sin k. (c- d) J WVc now numerically search, using Muller's method for example [.5, p.2621. to find the values of k9 which satisfy (41) and thus correspond to microstrip modes. With the propagation constants known, the fields in the ith layer can be easily found. These fields in the lavers are expressed in terms of the homogeneous solutions to Equations (1) and (2) which can be obtained by inspection of Equations (34 - 39). Their general forms for the it layer are = j(2 - 6m) EJ E- o 29rak.(k2 + k2) [i.U,NIee[k, k,(z - c)] - jIleo[k,, k(z - c)]] [c/LMee[k, kz(z - d)] - j Nleo[kx, k,(z -d)]] [7Uoo[k., k.(z - c)] + joe[k., k.(z - c)]] + Bi (44) [77mLNoo[kx, kz(z - d)] + j4oe[k,, kz(z - d)] and =- -s j(2- -6m)ki mO 2rak (k, 2 + k2) I ( [mUoo[k kz(z - c)] + jMoe[k, kz(z - c)]] 1 n [71mLMoo[kz kz(z - d)] + jloe[kz, kz(z -d)] 13

k(z- - C jNeo (-kr.k:(z - C) c [eu \ee[kt.(-d)]-jeo[kzs k( (-d)]] - } which are valid for all layers except the source layer. Some of the leading constants are preserved for convenience in later notation. The expansion of these expressions is given in Appendix A. From these, the power density can also be written and integrated analytically to determine the characteristic impedance as shown in Appendix B. IV. Numerical Results WVith these analytical results, the fields, power flow, characteristic impedance and propagation constants can be readily found for shielded strips with any combination of layered substrates and superstrates. Extensive comparisons with measurements available in the literature, other techniques, and a commercial CAD package (Touchstone [6]) have been made for the case of microstrip. In all instances, the accuracy of the technique has been shown to be excellent, including the ability to evaluate higher order modes. Some of the details are available in [7] but omitted here, since for the special case of microstrip the formulation could be greatly simplified. Of greater significance is the application to multilayer structures. MIultiple layers may be required by the circumstances, as we have pointed out, but also may be introduced to take advantage of their unique properties. To illustrate, let us examine by numerical experiment, the impedance and propagation behavior of the

structure sho'wn in Figuie 2. For convenience, let us assume that the dimensions are fixed except for the strip width. but we are at liberty to set the dielectric constants (cover height = a = 250 mils, I.. = 50. 1B = Ic = 2.5 mils). Furthermore. let us set the relative dielectric constant of laver B to CB = 13 as if the line were part of a circuit etched on GaAs. Observe the variation of the line impedance with frequency as a function of'c in Figure 3. In this figure the line width has been varied as well to achieve an 54.0.........I.......... 53.0 52.0 51.0 03 50.0 _ E 49,0 -.... = 348.0 _.a EC: = 13. W 16.9 mils - oL~ 47.0'~ 47.0 Ec = 8, W = 20.0 mils,~ 46.0 -. C =6 W =6 22.5 mils U 45.0. -.-.. -- = 4, W =27.3 mils - 44.0 C- -... e=2, W39.0mils 43.0... 0.0 2.0 4.0 6.0 8.0 10.0 12.0 14.0 16.0 18.0 20.0 Frequency (GHz) Figure 3: Impedance variation with frequency versus fc (CA = 2,:B = 13) impedance of 50Ql at f = 1.0 GHz. Thus, by selecting an appropriate value for 15

the sub-substrate permittivity. the variation of thie characteristic impedance.:: frequency can be minimized. This feature can be used to improve the design (oe broadlband highl-performance filters and matching networks in which the frequency response is sensitive to the variation of impedance. A similar effect is observed when the superstrate permittivitv is altered. From a different point of view, consider what happens when the line width is fixed. Figures 4 and 5 shows the variation of the impedance and effective relatiV-e dielectric constant as a function of the superstrate permittivity. The sub-substrate 65.0 I. ~A= 1 C: 60.0......-.. 2., CD a>~~~~~~~~~ —-— ^ = 4 55.0...... = 6 n:,,r,~~~~~... C=8 50.0 _ —.. _ _ _-_.- -.......: —---- 35.0 3 45.0 0.0 10.0 20.0 Frequency (GHz) Figure 4: Impedance variation with frequency versus EA (EB = 13, EC = 8, = 16.8 mils). 16

10.0 0- ~ E - - - - = 8 0.0 10.0 20.0 Frequency (GHz) Figure 5: Effective relative dielectric constant variation with frequency versus EA (cB = 13, Ec = 8. W = 16.8 mils). -~~1 4~~~~~1 17

Cielectrlic constant has been set to -c = S to flatten the impedance variatio,In - lrequcency as before. We note that the propagation constant for the line. inici:. ec 1)% Figure.5. displays the opposite sensitivity to superstrate permittivit. tiat the characteristic impedance becomes more stable as fA is raised but the propagation constant becomes more sensitive. This suggests a potential mechanism for post fabrication tuning of circuits where the frequency response is sensitive to electrical line lengths. For example, the resonant frequency of a tuning stub could be raised or lowered by changing the value of the superstrate permittivitv. relative to a predetermined value. If the nominal relative dielectric constant for the superstrate is high, the impedance of the stub would not be significantly altered. The superstrate could be applied after some preliminary measurements have determined the required value of dielectric constant. Besides the obvious utility of this type of information for circuit design, the ability to visualize the field or power distribution often provides important insights into why certain structures behave as they do, and also, how the behavior might change when the structure is modified. Examples include the plotting of field magnitudes on the cross section of stripline as shown in Figure 6; microstrip. Figure 7; and suspended microstrip, Figure 8. The upper part of the structure in Figure S has the same dimensions as the structure in Figure 7. By comparing the fields in these two cases, especially in the substrates, one can appreciate how one structure behaves quite differently from the other. Consider for example, what would happen if another strip is introduced on the substrate in these two cases.

E Field on waveguide cross section Arrow scaled to logarithm of field magnitude 0dB > -20 dB -40 dBH Field on waveguide cross section Figure 6: Stripline field distribution for the dominant propagating mode. Clearly, the suspended microstrip field distribution would be more greatly disturbed than the conventional shielded microstrip, therefore the coupling between strips would likewise be greater. Such an ability to visualize the field demonstrates one of the benefits of this type of analysis: enhancing our intuitive understanding of the behavior of these structures as well as providing quantitative information. Finally, the fields for the multilayer case discussed earlier are shown in Figure 9 with CA = 8, CB = 13, *c = 6. _~~~~~~~~~1

E Field on waveguide cross section H Field on waveguide cross section Arrow scaled to logarithm of field magnitude Arrow scaled to logarithm of field magnitude 0 dB - 0 dB-20 dB - -20 dB - -40 dB - -40 dB Fgr 7. d f - - - r I I I t _ _ _ I I I _- -' 9 I _ - Figure 7: Microstrip field distribution for the dominant propagating mode. 20

E Field on waveguide cross section H Field on waveguide cross section Arrow scaled to logarithm of field magnitude Arrow scaled to logarithm of field magnitude 0 dB - 0 dB - -20 dB -20 dB -40 dB - -40 dB - ~'" ~' \ f!,, ~' -- A l N, I 1 0,, ". -, =n ~~ t / f V \ \j — _ _,' s/! I I!_ X _ _ - \ \ t t _ of / # # \ s _ _ X X s _J t l - I *. I t I I! t I Figure 8: Suspended microstrip field distribution for the dominant propagating mode. 21

E Field on waveguide cross section H Field on waveguide cross section Arrow scaled to logarithm of field magnitude Arrow scaled to logarithm of field magnitude 0 dB -- 0 dB -- -20dB- -20dB -40 dB- -40 dB \ * % 9 9 I& I -,,, - %, I I I I' %.', I I J I ~ s -' _ \ S _W * _ _ d \ _ S _"* _ _ 9, - J, _ _, I I - _' * _.-. _ \ t / - I i I / - - _ \ \ / _ ^ I -,, / \ t t f t Figure 9: Fields in a multilayer structure for the dominawnt propagating mode. 99~~~~~~~~~~-:!!i!''!/-:~:~ "' -......'........'' -..:..:.!~..~.:.:...:.:.:Z:.:.:.:.:...:~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~:;i~::~~::~

Appendices A. Expansion of Fields LSE.NIo,l-s Expanding the V'WAFs we find that the LSE mode fields in each layer can be writtcn in the form ELSE = A ki( [Zijk-:ky cos k,.x sin k:x' - yk2 sin k,x sin k'] cos 1 J,[{ cos k,,(c - z) [u,v + j tan k,,(c4 - z)] (i < O) cos k.. ( ci -'d) cos k, (Z - di) [i,Li - j tan ki (z - di)] (i > 0) (A.1) LHLSE; =:FA, [a(k2 + 2) [ sin kxz sin kxz' + yjkkky cos kzx sin kzx'] cos 1 fk,ic cos k,i(ci - z) [uv;i tan k,i(c - z) - j] (i < 0)'cosk'(ci -di) 1 cos ki(z - di) [ieLi tan k,i(z - di) + j] (i > 0) +Ai, [a ki -[] k - cos k4x sin khx'] 1os k,i( d{ cos k,i(ci - z) [7,c; + j tan k,i(ci - z) (i'cos,(:-d') cos k,i(z - di) [O.L - j tan k,(z - di)] (i > 0) (A.2) where i = 0 for the layer containing the strip and -r/eLO Ao = (i'UO- aeLO) COS k(Co- do) - d j(7UO7eLo'- 1)sin kZ(co - do) By matching the tangential components of the field at the interfaces above and below the strip layer, the remaining coupling coefficients are found to be ~Al =Ao I' [uo + j tan kzo(co - do) (A.4) kz = Ao tan k (c-dl(A).4) k41 L AL1 — jtank.l(cl - dr)

AI. —i k:,'eL( -I) /= t',- cos /-k.(,_(c,_ 4 - di_l )eL, - j tan kt,(c, - d, )! 4 = *4 +c kz: T`eLTh11 Ik I,-1) Cos k:(,~l)(Ci+l - d,+l )F[/.' - j tan kA:(c, - d, )0 LSNI N[odes Similarlyv. the LSMI mode fields can be written as ELS,, = -Bi [a;2j2 + )] [ikPiky;k cos kz x sin ks:' + yyk2 sin k:z sin k:x'] c 1~, fc; cos ki(ci- z) [imLi + j tan k:i(c - )] (i O) c - d cos k,i(z - d) [Im - j tan k,j(z - di)] (i > 0) iB,[ 2 - [),z0jky sin k,: sin k,x'] c 1k, (; ){ cos kzi(ci - z) [,mui tan k,i(ci - z) - j] ( O) cos k.,(c, - d,) cos k~i(z- di) [r1mLi tan k,i(z - d,) + j] (i > O) (A.6) HLsl ~ = rs ~c k~+ k2 2 [HL'M = 2j k)] [-k sin kx sin kz'- j kk cos kx sin k x'] cos 1 fc; cos k,i(ci - z) [.mui tan kzi(ci - z) - j] (i < O) cos k,i(c4-di) | cos k.i(z - d) [1mLi tan k,(z - di) + j (Z > 0) (A.7) where the coefficients are defined by — mLO B= (muVOm- OmLO) COS kz(co - do) -j(,mUO77mLO 1) sin k(co - do) (A. B = Bookk2 [r/mUo +'jtan k.o(o -do)] (A.9) Bokzl i2 l - j tan kzl (cl - dl) 24

, kzl cos k(,-_l)(c,-l - d,_,)[~imLi - j tan k, (c, - d, )j B,+1 k..(,:+l) k2m,L(i ) (, k,k(+) cos kz(,+l)(c,+l - d,+)[Oe,t, + j tank. i(c - d, )] I A.!O' To calculate the field at a particular point in the ith laver. we then calculate only the coupling coefficients (Ao... Ai) and (Bo... Bi) while evaluating Equations (A.1.A.2) and (A.6,A.7). B. Power Densitv Integration Calculation of Power For computation of power (P), the needed component of Poynting vector is Y * E x H. For the case of LSE modes, power density is given by 2wP H [ 4 ] [ATkr] cos2k,x[f sin k,']2 ~P= E,H; k Ik, (kdk2 + k2) ki Icos k,2(c - di){2 0,ucri cos kzi(ci - z) + j sin k,i(ci - z)12 (i < O) {ilLi cos kzi(z - di) -sin k,i(z - di) (i > 0) (B.1) For the LS%M modes the required term for calculating power density is P-E = -E2Hw'.i [B,k?][ B'(k;)2 1 sin2 k,,x [f sink.,x']2 z 2 a2 [ J](k2 + k2) J cos k,(ci - d,)l2 {Imui sin k,i(ci - z) - j cos k,i(c _- z)12 (i < 0) {ImLi sin k,i(z -di) + j cos kzi(z - di) (i > 0) (B.2) 25

Tllere are also two sets of cross-terms from the ELS.A x HLSE product neede-i to calculate powver density. They are 2... F B,kkykzk,, i A rk: cos2 krx sin krx12? = 2,HI =- a2 Lk2(k2+k k2) k; cos kC, (c - dd,)j2 { [Ln,cos, kC (c, - z) + j sin k,(c- )] cos,c,- z) + j sin k, (c, - z)' [iimLi COS ki(z- d) - j sin k,i(z - di)] [iL cos k,(- d )- j sin k,i( — d, )i 2wi [B,k,j A]( k. 1 sin2 k,x [f sin k,x']2 2wE H B*=, y..2 ~P = - E2H- 3aZ [ k (k[ + k2+)J Icosk,j(c. -d,)l2 { [7lmui sin kz,i(c, - z) - j cos kjiz(ci - z)] ['ieUi sin k,i(ci - z) j cos k<i(c -' < 0 [rmLi sin k,i(z - di) + j cos k,i(z- di)] [ir/Li sin k-1i(z - d) + j cos k,i(z - d)]' i > 0) tB.4 The evaluation of total power can be done analytically by integrating the power density on the waveguide cross-section. Integration of the x dependence is trivial and has been indicated in the previous expressions. The z dependence appears in two forms, the first of which is c [rI cos k(c- z) + j sin k(c- )][2 cos k(c- z) + j sin k(c - z)]..r = / 1 [?1 cos k(z - d) - j sin k(z - d)][r72 cos k(z - d) - j sin k(z - d)] 77177; - 177 - 77' = 14Re (k) sins22e(k)(c- d):t: jr2R ( sin 3e(k)(c- d) 4~J?e(k) 2~e(k) + q177; + r sinh2 In m(k)(c-d) (B.D) 4Qm(k) s2( m(k) where Re(k) and Sm(k) symbolize the real and imaginary parts of k respectivel. The second form is c [71h sin k(c - z) - j cos k(c - z)]1[r2 sin k(c - z) -j cos k(c - z)]' IPN = dz [71 sin k(z - d) + j cos k(z - d)][r2 sin k(z - d) + j cos k(z - d)]' 26

=?'- " _ 2 sin 2.e(k)(c- d) sn e csinh2 1 k)sinh h2n(k)(c-d) r nk)( - d) iB.61 W ith these expressions. calculation of the characteristic impedance reduces to a simple summation of terms, once the matching coefficients are known for each layer and mode. 27

References [1] C. T. Tai, Dyadic Green's Functions in Electromagnetic Theory, International Textbook Company, Scranton, Pennsylvania, 1971. [2] C. T. Tai, "Dyadic Green's function for a rectangular waveguide filled with two dielectrics.", Journal of Electromagnetic Waves and Applications, Vol. 2, No. 3/4, 1988, pp. 245-253. [3] C. T. Tai, "Dyadic Green's functions for a coaxial line," IEEE Transactions on Antennas and Propagation, vol. AP-31, pp. 355-358, March 1983. [4] T. E. Vandeventer and P. B. Katehi, "An integral equation method for the evaluation of conductor and dielectric losses in high-frequency interconnects", IEEE Transactions on Microwave Theory and Techniques, Vol. MTT-37, No. 12, December 1989, pp. 1964-1972. [5] W. H. Press, B. P. Flannery, S. A. Teukolsky and W. T. Vetterling, Numerical Recipes, Cambridge University Press, Cambridge, 1986. [6] EESOF Inc., Westlake Village, CA. 91362. [7] N. L. VandenBerg, "Full-wave Analysis of Microstrip-fed Slot Antennas and Couplers," Ph.D. dissertation, The University of Michigan, 1991.

APPENDIX D Norman L.VandenBerg Investigation of a Strip-Fed Doubled-Tuned Slot Coupling Through Waveguide Sections August 6, 1990

Investigation of Strip-fed Double-tuned Slot Coupling through Waveguide Sections N. L. VandenBerg and P. B. Katehi It has been suggested that it may be possible to increase the operating bandwidth of a radiating slot such as the one illustrated in Figure 1, by adjusting the slots on the ends of the waveguide section to different lengths. This would'stagger tune' each slot to a different frequenzy which should increase the overall bandwidth. Radiating Slot Rectangular Coupling i. Waveguide Slot Stripline Figure 1: Stripline excitation of a waveguide-fed slot. Since actual dimensions are not yet available, we have originated a set of dimensions for a specific study case to investigate the feasibility of this approach. As previously proposed, we have modified and augmented existing programs to handle the intervening waveguide section which can now have different dimensions than the end slots ( - differing from our previous report on thick slots). The case to be studied is illustrated as a wire-line drawing in Figure 2.

l-stLor Figure 2: Wire-line drawing of the study case geometry. Before proceeding to the results, observe that the addition of the waveguide section introduces many new parameters including: * Waveguide length. * Slot width to waveguide height ratio. * Slot length to waveguide width ratio. * Relative position of slot in waveguide width and height dimensions. The parameters for the results to follow have not been chosen to optimize the performance in any way other than the reasoning that will be given. In fact, at this point, we don't completely know how the characteristics will vary as a function of these parameters. A study of these effects will be performed at a later date. To start, we assumed 9.7 GHz (X-band) as the target center frequency. We also assume for this case, a cavity on the order of A0/2 in size using Duriod ( c, = 2.2 ) fed by stripline with thickness of.125" (sandwich of.0625" boards). Based on the results of our previous report for thick slots, we expect that the addition of

a length of waveguide may increase the resonant length of the slot. Therefore, initially we should find a case with the resonant slot length of a'thin slot' short enough to provide a margin for a later increase in length. A cavity 0.60" square and a dielectric cover ( fr = 2.2, t =.0625 "), produces a resonant length of around La = 0.40" for a.10" wide slot. A strip width of 0.10" was used to feed the slot which is approximately a 502 line. The behavior of the slot as a function of slot length is illustrated in Figures 3 - 6. Since the behavior with respect to length is related to the frequency response through its relationship to wavelength, these are an indication of the frequency behavior of the slot as well. Using length as a parameter is preferred because this data is more easily obtained than direct frequency data. More will be said about this later. Figure 5 and 6 suggest that the zero reactance length may not be the best definition for resonant length. One can see that the curves come close to an inflection point at a non-zero reactance of ~ 0.15Q's. If we were to add some capacitance to offset the reactance to the inflection point, we could conceivably get a zero-reactance length which is virtually constant with offset. Or we could find a point where the zero-reactance point comes closer to matching the slope of the peak-resistance curve. We will see later that with the addition of a waveguide section the reactance offset gets even larger, to the point where the reactance no longer crosses zero without compensation. For the moment, we will accept the peak-resistance point as the resonant length. Wte now introduce a section of waveguide between the slot in the half-space

("top slot") and the slot in the cavity ("bottom slot"). For significant coupling to the radiating slot for a waveguide of substantial length, the waveguide must be operating above the cutoff frequency. Thus we chose the waveguide to be halfheight X-band waveguide with transverse dimensions of.900 " x.225 ". The results for the same cavity with the waveguide section added are given in Figures 7 and 8. In these figures, the resistance and reactance is given as a function of slot length and waveguide length. For all cases the slots on either end are identical in length. As you can see, the waveguide has the affect of shortening the resonant length of the slots as the waveguide length is increased from the initial value of Lw =.2 " to a value between.7 and.8 ". At Lw =.8 " it appears that the resonant length has jumped to a much longer value (outside the range of the graph). The curve for Lw =.9" suggests that the cycle will probably repeat as the waveguide is made progressively longer. The reactance curves illustrate the point mentioned earlier since for certain values of LW there is no zero crossover point. Note that over a certain range of values, the affect of increasing LW on the peak value of resistance is qualitatively the same to that of increasing the offset of the stripline for the'thin slot' case in Figure 3. Also, the affect on the apparent bandwidth of the slot is the same, that is, as the waveguide length - or offset for the'thin slot' case - is increased, the bandwidth decreases. Thus we have already, a'rule' for increased bandwidth: minimize the waveguide length and/or the strip offset. Of course, to achieve a given slot impedance may then require (broadband)

impedance transformers which may not be a desirable or practical option. The affect of detuning the slots can be seen in Figures 9 and 10. These curves are based on the arbitrarily selected case where the waveguide length is fixed at Lw =.5 " with a fixed offset of s =.0138 ". Note that here the coupling (resistance) decreases as the difference in slot lengths is increased and accordingly the apparent bandwidth increases. Again this can be compensated for by external impedance transformers or by simultaneously adjusting the offset and/or waveguide length. To estimate the improvement in bandwidth, several cases were run while attempting to compensate for the change in resistance. The resistance curves for three cases are shown in Figure 11. In these cases, the waveguide length is fixed at Lw =.5 ". The difference in slot lengths is varied while simultaneously changing the strip offset in order to achieve approximately a constant resistance. (An alternative would have been to vary the waveguide length as pointed out earlier. It is not yet known whether either method should be preferred over the other). The normalized resistance curves are then re-normalized to the peak value to remove any remaining variations. Slot lengths are also normalized to the resonant lengths to remove variations. An increase in bandwidth is now obvious. To estimate the bandwidth improvement we arbitrarily define bandwidth by the 1/IV points on the resistance curve. The true bandwidth would depend on the external circuits which are actually used to feed and load the slot. The adequacy of the definition also depends on how well the'versus length' data translates into'versus frequency' which will be discussed later on. Assuming this definition of

bandwidth can be applied, the bandwidth of the case with equal length slots is approximately 4.5 %. For the widest case it is about 6.5 % representing an increase of approximately 45 %. Considering the limited exploration of the parameters used so far, it is likely that greater increases could be achieved. Now because each transition in the structure may not have the same response variation with frequency, to rigorously translate the previous figures into a frequency response, it is necessary to run each geometry at an ensemble of frequencies. An alternative would be to use a transformation based on a first order approximation which assumes that all of the transitions in the frequency domain behave as 1/ length. Then the length data can be transformed analytically by the relation: Af AL fo Lo However in our case, since we are at resonance, the assumption is poor and the transformation does not produce frequency response results which compare well with the actual frequency calculations, although it may be adequate for resonant length/frequency shifts. Therefore for true frequency response results we must repeat the calculations at each frequency point. This has been done and the results are shown in Figure 12. Based on the same definitions, this produces an even greater increase in apparent bandwidth, in the neighborhood of 55 %. Hence, since the previous estimate based on length calculations was conservative, we may be justified in our preference for the lengthbased results. The reason for this preference is because it is much more costly to generate different frequency points than it is to vary the geometry.

In conclusion it has been shown that, based on the theoretical model we have developed, the concept of double-tuned waveguide coupled slots is a feasible approach for increasing bandwidth. Within the lirrmited range of parameter variations used here, we have shown bandwidth increases on the order of 50 %. It seems likely that with a more detailed study of influence of various parameters on the impedance characteristics, even greater improvements in bandwidth could be obtained. From a practical standpoint, the work done so far has raised a number of questions which should be answered to guide the scope of the theoretical study so that it will be in-line with later experimental effort: 1. Do manufacturing considerations provide any restrictions or limits on any of the parameters (dimensions) above or others not yet covered? 2. For the above, we have assumed a cavity-backed slot with cavity dimensions on the order of A/2 with a two-dimensional array in mind. Does this fit in with your intended approach at all? What is your current concept for the stripline or microstrip feed network and the arrangement of the array? (The data above is for a two-port element). 3. What are the requirements for the impedance levels of the slots? Are impedance transformers acceptable? How (where) would they be arranged? 4. What are the bandwidth requirements and how should it be defined if different than above? 5. What and where are dielectric materials to be used or allowed?

In the interest of efficiency, it would be probably be appropriate to address these issues as they relate to your antenna concept and experimental work before the theoretical work goes too far based on assumptions. Ideally, the model based characterization will be performed on realistic geometries which can later be compared to your experimental measurements.

1.80 1.60' A1.40 tO. \ s.0266 nb,m 1.20 _,O:~ 01.00 0-3 A E5 0/ # A. ~ —... ~. 11 @: 0 X""<~~~~~~" " "'-:-.0032 ce 0.60 --'.,:r ~ ~ ~ ~ ~ ~ ~ ~ a -. -.0074 indu # * / ~ ~ me.. *%~ % ~ 0.40 [ - /..."./'.-..... "-' -.0117 "'...'.-.oll7indi 0.0 S*.0160 incb - —.0202 ind 0.00.... 0.33 0.34 0.35 0.36 0.37 0.38 0.39 0.40 OA1 0.42 OA3 0.44 OA5 Slot Length (inches) Figure 3: Normalized slot resistance as a function of slot length for various stripline offsets (s). Slot is assumed to be negligibly thin and covered by a dielectric slab on an infinite ground plane.

1.10 0.90..0266 i c.h ~- s=X0S3 0.70,......; - 18... ~ 0.30 sw* h- - "'- ] 0~310'.0053 inches -.0.10 s \.,';;'- I=-. ol7inches -0.30 s - -.0.... s-.0inches'\.'-.::.....0 s0 - -.02 inced 033 0.34 035 0.36 0.37 038 0.39 0.40 0.41 0.42 0.43 0.44 OA5 Slot Length (inches) Figure 4: Normalized slot resistance versus offset. 10

2.00............1............ I..''........'''I'.. 1.80 1.60 1.40 1.20 o 1i.00 _ _ -' 0.80 0.60 o 0.40 0.20 0.00.............................. -0.025 -0.015 -0.005 0.005 0.015 0.025 Offset (inches) Figure 5: Normalized slot reactance as a function of slot length for various stripline offsets (s). 11

0.425.... 0.420, 0.415 Zeo Reactance Legth.... Peak Rcsistance Length 0.410 0.405 0.400 ------.. 0' 0.395 0.390 0.385......... I.... -.020 -.016 -.011 -.006 -.001 0.003 0.008 0.013 0.017 0.022 0.027 Offset (inches) Figure 6: Resonant slot length versus offset. 12

1.20' ci 1.00 Lw 2 d2 iwlm l l......... ic s 9 0.80 ------ LW=.4 inc.a 0.60 LW;.6 inchs ~ LW.9 irrch 0.20 I 0 x I ".......... Slot Length (inches) Figure 7: Normalized slot resistance versus waveguide length ( Lw). Both slots are of equal length with strip offset fixed at.0138 ". 13

1.10 0.90,:N ] g o'Ii Llw:2 I bwls Ai ~ ~ ~ ~~ ~ ~~ ~~~~~~....... =.5 ks -0.10 /.'" -0.30 - 0.320 0340 0360 0380 0.400 0A20 0.440 0.460 Slot Length (inches) Figure 8: Normalized slot reactance versus waveguide length ( Lw). Both slots are of equal length with strip offset fixed at.0138". 14

0.50 0.45 a, 0.40 - - t. a S mils 0.30...... A - 12.8 mils 0 0.25.... - 17.0 mils 0.20..... &,- 213 mils'~ 0.15 ~ 0.15 - a-25.Smils Z3 ~ r a = 29.8 mils 0.10 = 34.0 mils a 38.3 mils 0.00 0.35 0.36 0.37 0.38 0.39 0.40 0.41 0.42 0A3 0.44 0.45 0.46 0.47 Lower Slot Length (inches) Figure 9: Normalized slot resistance as a function of lower (cavity) slot length with the difference in slot lengths as a parameter. 15

0.65 I I. 0.60. 0.55 -. u 035 ~ \- 0.0 milo.'A........ 174.3 zui o_so 0.3 ---—. Aw13miL 0.30 ~~~~~~~ ~~~ ea~~~=s.sm,i 0.25 a 29.85 mila 0.45, \ 0. 4 0. 0.38 0 O 04.2404.5.46 0 Lowe.r So Lg17.0 mih C4 035 ~~i r1m 0.30 - -, rsondnc3 o Fiur 9- x;\ \ 16'~~~ --- d...A=25.5 mils 0.25 -------- a - 29.8 mils 0.20 - q-="~::':::"~ ~ ~ ~ n340 il 0.15 0.10r~C'T~-~~~,~ C A =. 38.3 mils 0.10 0.35 0.36 037 0.38 0.39 0.40 0.41 0.42 OA3 0.44 OA5 0.46 OA7 Lower Slot Length (inches) Figure 10: Normalized slot reactance for vaxious differences in slot lengths. Correspondence to Figure 9 16

1.00.-.. cc 0.90''~.~~~~~~~~~~~~~~I / // 0.70. I0\. 0850 C.) I, 0.40r v, 1: -, *n 0.70 -~ o 02 \',', IL I" " 0.50, I. 0.40 I, "t'-]i 0.30 rI5 0.92 0.94 0.96 0.98 1.00 1.02 1.04 1.06 1.08 1.10 1.12 1.14 1.16 Normalized Slot Length L. I LS Figure 11: Compensated normalized-normalized slot rt.038 incesistance vs. normalized slot00 0.20l e Z~~ j /jj..... offtt=-.os21 in~i~s, a = 3s.3 muJi 0.92 0.94 0.96 0.98 1.00 1.02 1.04 1.06 1.08 1.10 1.12 1.14 1.16 Normalized Slot Length Ls / L~,~ length. 17

0.50 0.45 0.40 C: 0.35 0.30 0.25 0.20 0.15 Frequency (GHz) Figure 12: Frequency response: normalized-normalized slot resistance vs. frequency. 18

APPENDIX E Norman L.VandenBerg Full-Wave Analysis of Aperture Coupled Shielded Microstrip Lines October 15, 1990

Full-wave Analysis of Aperture Coupled Shielded Microstrip Lines' N. L. VandenBerg and P. B. Katehi Radiation Laboratory, The University of M.ichigan, Ann Arbor. II Abstract A full-wave space-domain integral equation analysis of aperture coupled shielded microstrip lines is presented based on Pocklington's integrals and the equivalence principle. The derivation of the associated dyadic Green's functions in the form of waveguide LSE and LSM modes is described. The line currents and slot voltage are expanded in terms of subsectional basis functions and the method of moments, together with even and odd mode transmission line analysis, is applied to determine the two-port scattering parameters. A particular case is illustrated which demonstrates the behavior of the coupler as a bandpass interconnect. Introduction Application of monolithic techniques to microwave and millimeter-wave circuits has introduced the need for accurate full-wave analysis to account for the effects of the shielding structure and also the interaction of circuit elements which may be closely spaced. These effects may not be accounted for by other methods such as simple transmission line analysis, especially as frequency increases. Additionally, high-frequency interconnect approaches are required which must be accurately modeled for design and also must fit well in a monolithic fabrication scheme. Transitions from microstrip to slotline lhave long been recognized as impolrtaiit circuit elements. Numerous investigators have presented approximate analytical techniques to characterize these structures with applications to circuit elements [1] - [4]. A full-wave analysis for this transition was reported in [5] with applications to open structures. 1This paper is a revision of the original which appeared in tile 1990 Microwave Theory and Techniques Sym.lposium Digest reflecting changes in the Numerical Results section after the computer program was modified to improve accuracy.

In this paper, the analysis of aperture coupled shielded microstrip lines is presented. This structure, in a variety of forms, has a wide range of applications to both broadband and narrowband connections and can be used as a building block for interconnects [6], phase shifters and inverters [7]. directional couplers [S' as well as filters. A quasi-static analysis has been provided in [9], however. this may not be sufficient, particularly for higher frequencies where end effects and highei older mode coupling become more significant. A more recent paper [10] presents a transmission line analysis with excellent results, however, similar shortcomings would be expected. Hybrid methods which combine two-dimensional full-wave analysis with transmission line theory, as in [11], should certainly extend the validity of such models, however, may still not account for all discontinuity effects. Our approach uses a three-dimensional full-wave space-domain integral equation method employing the method of moments with Galerkin's procedure to account for all possible interactions. Analysis The basic structure of the coupler to be discussed is as shown in Figure 1. Variations on this geometry include cases where the strips are on the same substrate in a single cavity, addition of multiple layers to the substrates/superstrates, reverse couplers where the lines exit on the same wall, parallel slots and lines, among others, but can all be analyzed using the same approach. The analysis proceeds as follows: the slot is replaced on both sides by an equivalent magnetic current - representing the tangential electric field in the slot - backed by a perfectly conducting wall (all walls will be assumed to be perfect conductors). In this case, the problem is now separated into two independent regions, coupled together by the magnetic current. Using the same current on either side enforces the continuity of the electric field in the slot. The fields in the cavities can now be written in term's of Pocklington integrals as follows: E =-jwI J GeJ J dS'-< GeK K dS' (1) H = JGmJ 3 dS' -j4 jJJGmK Ix dS' (2) The subscripts on the Green's functions indicate whether the function is of thle electric (e) or magnetic (m) field type for electric (J) or magnetic (K) currents. By using the appropriate rectangular cavity Green's functions, the fields in thle cavities satisfy the boundary conditions on the walls. The:,2maining boundary conditions, zero tangential electric field on the strips and continuous tangential magnetic field in the slot, allow us to write the integral equations which are -j tri p,L dS-,K I= (3) 2V I,,it-n

]ItimJ' JL dS-jw [ELGmK + eUGmKJ I~dS trIPL ~3L Iid"lot =slot - - ]1 Gmj JU dS'= 0 (4) trTip where 0' implies that the field is non-zero at gap generator locations. The superscript indicates the evaluation point for the field. The Green's functions are solutions to the dyadic equations: V x V x Gej - k2GeJ = I(R-') (6) V x V x mJ- k2~mJ = V x [IT(R-')] (7) V x V x GmK - k2Gm = I6(R - R') (8) V x V x GeK -ck2GeK = V x I6(R- R') (9) Solutions to these equations can be found for geometries of the type illustrated in Figure 2 where the impedance boundary conditions allow for treatment of multilayered substrates and superstrates as may be encountered in monolithic circuits. The Green's function is found for the layer containing tlhe strips or slot by applying transmission line theory to the other layers to obtain the impedance boundary conditions (,q) corresponding to each mode. Subsequently, the fields in the other layers can be found from the homogeneous solutions to Equations (6 - 9) and a pair of coupling coefficients for each layer. The solutions are expanded in terms of LSE and LSM modes which facilitates finding the fields in the remaining layers since these modes are decoupled on the boundaries and can be individually matched to like modes in adjacent layers. The coupling coefficients are then readily found by matching the tangential field components at successive layers. The longitudinal current components are now expanded in terms of piecewise sinusoidal functions with a hMaxwellian transverse distribution which satisfies the edge conditions. In general, the currents are written in terms of bothl longitudinal and transverse components however, in this case, it will be assumed that bothl the slot and the strips are narrow enough so that the longitudinal components of current dominate their behavior and the transverse components can be neglected. The integral equations can now be written as a matrix equation in conventional method of moments fashion. The inverted matrix is then used with even and odd gap generator excitations at the line ends to find the currents on the microstrip lines. From the even anlcl odd currents, the scattering parameters can be found which characterize the coul)ling behavior.

Numerical Results A coupler with the geometry of Figure 1 was analyzed using the above tecilniques. The parameters whicll can be varied in this design are numerous. coilsequently, only a few variations will be presented here. In all cases, altliough not required in general, symmetric geometry is maintained to simplifv the even and odd mode analysis. The affect of the slot length is shown in Figure 3. It can be seen that the slot is initially too long for an ideal match at the frequency shown ( 11 GHz ). As the slot is progressively shortened, at a certain length the slot "matches" the two port coupler and with further shortening the match gets progressively worse. Using a transmission line analogy, one can interpret this effect by transforming the impedances at the ends of the slot to the center. These end impedances are nearly short circuits, the difference being due to fringing fields which extend beyond the ends of the slot line, fully accounted for by the full-wave analysis. At the matching length, the resulting transformed reactance at the center, cancels the reactance associated with the junction thus matching the two ports. As the slot becomes very short, the field in the slot is effectively'"siort circuited", thus coupling is reduced. Thus S21 tends to zero while S11 approaches unity (since the structure is closed and assumed lossless). The phase of S11 referenced to the end of the line then becomes the phase due to the end effect, typically, -10 to -15 degrees. All of these effects would be expected to repeat as the slot lenlgtll increases in multiples of A, however, for the case studied here, the maxinlum slot length is limited by the dimensions of the shielding package (.25 x.25 inches) whlicl are chosen to allow only the dominant microstrip mode to propagate. Figure 4 illustrates the same phenomena in the frequency domain. By sllortening the length of the slot, we are actually moving the center of the frequency response. The shape of the frequency response does not remain constant witll tile shift however, because tile high end is dominated by a null due to the line stub. This affect appears at the frequency for which the line stub is approximately A/'2 in length so that there is a virtual open circuit beneath the slot, in whichl case there is very little coupling between the line and slot. Figure 5 illustrates this affect by showing the influence of the line stub length (1 - see Figure 1)on the frequency response. Variation of these stub lengths has the effect of changing the position of the current maxima (virtual shorts) and minima (virtual opens) on the lines relative to the slot, thus varying the degree of coupling through the slot. Consequently, the peak in the frequency response roughly corresponds to a line stub length which places a current maximum below the slot (~- A/4). In Figure 6 the affect of the line separation parameter (s - Figure 1) on tlhe phase response is shown. Note the increase in phase shift due to tile added lcngtll of slotline as the separation is increased. Also notice that for no separation, as the line stub lengths approach zero, the phase shift is near 180 degrees (the reference

planes are set at the location of the slot). Thlis additional phase shift is due to geometry as has been pointed out in [7] and could be taken advantage of in combination with other slots for phase shifters. The frequency response shown in Figure 7 demonstrates the utility of tlle structure as a filter. NWNith proper selection of the geometric parameters such as line and slot widths, line separation and substrate heights, the frequency response can be tailored to give the required center frequency, bandwidth, shape, etc. As can be seen, separation of the transitions enhances the bandwidth of the coupler to better than octave performance, similar to that which was demonstrated in [10]. In all of the cases presented here, the width of the slot was the same as the 1-ine width L.02 inches on c, = 9.7, lV/h = 1 substrate) resulting in a microstrip-to-slotline characteristic impedance mismatch of about 2: l. Matching the slot impedance and modifying the width of the stubs as in [10] to alter the frequency response cani also be readily accomplished with this method. Unfortunately, because the slot length for this simple geometry is typically on the order of A/2 to get low loss coupling, it may not represent the best choice for monolithic integrated circuit applications where small size is required. Therefore, future analytical and experimental efforts are planned to study and optimnize the performance/size tradeoffs for this class of couplers while taking full advantage of the benlefits of full-wave analysis for high frequency applications. For example, it may be possible to introduce combinations of shorter slots of \arving lengtlhs to produce a broadband frequency response whlile maintainiing a reasonably small area of chip real estate. Conclusions A set of integral equations for aperture coupled shielded microstrip lines has been introduced based on Pocklington's integrals and the equivalence principle. The associated dyadic Green's functions in the form of waveguide LSE and LSMI modes have been derived which allow for a full-wave analysis, accounting for all electromagnetic interactions of a microstrip - slot coupler. By expanding the unknown line currents and slot voltage in terms of subsectional basis functions and applying the method of moments together with even and odd mode transmission line analysis, the two-port scattering coefficients can be determined. The method'has been applied to a particular case which demonstrates the behavior of the coupler as a bandpass interconnect. Acknowledgement This work was supported by the National Science Foundation under contract ECS:8657951 and partially by the NASA Center for Space Terahertz Technology.

References [1j S. B. Cohn, "Slot line on a dielectric substrate." IEEE Transactions on.lIicrowave Theory and Techniques, \'ol. MITT-17, No. 10, October 1969. pp. 768-778. [2] E. A. Mariani, C. P. Heinzman, J. P. Agrios and S. B. Cohn, "Slot line characteristics." IEEE Transactions on Mlicrowave Theory and Techniqu ct. NVol. MITT-17, No. 12, December 1969, pp. 1091-1096. [3] E. A. MIariani and J. P. Agrios, "Slot-line filters and couplers." IEEE Trallsactions on Microwave Theory and Techniques, Vol. MITT-18, No. 12, December 1970, pp. 1089-1095. [4] J. B. Knorr, "Slot-line transitions." IEEE Transactions on M1icrowave Theory and Techniques, Vol. MTT-22, No. 5, May 1974, pp. 548-554. [5] H. Yang and N. G. Alexopoulos, "A dynamic model for microstrip-slotline transition and related structures." IEEE Transactions on Microwave Tlheo ry and Techniques, Vol. MTT-36, No. 2, February 1988, pp. 286-293. [6] R. H. Jansen, R. G. Arnold and I. G. Eddison, "A comprehensive CAD approach to the design of MMIC's up to hIM-wave frequencies." IEEE Tr-ansactions on Microwave Theory and Techniques, Vol. MITT-36, No. 2, Febrluary 1988, pp. 208-219. [7] I1. G. Gupta, R. Garg and I. J. Bahl, Microstrip Lines and Slotlines, Artech House, Dedham, MA, 1979. [8] T. Tanaka, K. Tsunoda and M Aikawa, "Slot-coupled directional couplers between double-sided substrate microstrip lines and their applications." IEEE Transactions on Microwave Theory and Techniques, Vol. MTT-36, No. 12. December 1988, pp. 1752-1757. [9] S. Yamamoto, T. Azakami and K. Itakura, "Slit-coupled strip transnmission lines." IEEE Transactions on Microwave Theory and Techniques, Vol. MITT14, No. 11, November 1966, pp. 542-553. [10] B. Schuippert, "Microstrip/slotline transitions: hModeling and experimental investigation." IEEE Transactions on AMicrowave Theory and Techniques, Vol. MTT-36, No. 8, August 1988, pp. 1272-1282. [11] H. Ogawa, T. Hirota and M. Aikawa, "New MIC power dividers using coupled microstrip-slot lines: Two-sided MIC power dividers." IEEE Transactions on Microwave Theory and Techniques, Vol. MTT-33, No. 11, November 1985, pp.1 155-1164.

i v-s 2c'' v /,.O'. K —--— ~ ---- IIu/.-.... *....k'..*, *rl ~ K i Figure 2: Sample multi-layered structure.

-3.00 P' -6.00:.) -9.00 -12.00 -18.00 -21.00..... i.... I..... l.......!....l...,.... l...... 0.000 0.025 0.050 0.075 0.100 0.125 0.150 0.175 0.200 0.225 0.250 Slot Length (inches) Figure 3: Affect of slot length on S21 and Sll magnitudes (f = 11.0 GlIz ). -6.0 7 -21/.0,,,, X \...', -9.0 ~ -12.0 -15.0, Slot LAaStb: 0.2500 inchea..oi,.....,SlotL.gtb 0.132inche -21.0 4.0 6.0 8.0 10.0 12.0 14.0 16.0 Frequency (GHz) Figure 4: Coupler frequency response variation with slot length. ~~P r t~~~~~

, -12.0' 18. 00:"/.. u Line Stub. 0.1250 inches -15.00 90/.00' -—. Line Stub: 0.0815 mchs -04.00 Line Stub: 0.0380 iches -21.0 0 - --- 4.0 6.0 8.0 10.0 12.0 14.0 16.0 18.0 20.0 Frequency (GHz) Figure 5: Coupler frequency response variation with line stub length. 180.00,................ 135.00 - E 0.00 -10.0.. 40.0025 O.O0 0.075 0.100 0.125 - 0. 0..-eLine Stub 0.Lnsl0 inchcs) Figure 6: Variation of S21 phase with line stub lengths ( f = 11.0 GHz ).

0.0 1 —&zwe 6, A- 9 0p - -3.0 c -3.o0- /.6.0. I I.', rr -9.0 6 1 a ao -12.0 d ~ ~ ~,'. Figure 7: Coupler frequency response variation with line stub length ( s =.0625 inches........I..... Lau........I.....10 ~21.0 -".. 4.0 6.0 8.0 1 0.0 12.0 14.0 16.0 18.0 20.0 Frequency (GHz) Figure 7: Coupler frequency response variation with line stub length (s =.0625 inches ). 10

APPENDIX F Norman L.VandenBerg Proposed Dimensions for Cavity-Backed Fixture Measurements October 24, 1990

Proposed Dimensions for Cavity-backed Fixture _Measurements (October 24, 1990) In our discussions on October 11 at TI. we offered to generate some numerical data which would serve as guidelines for the dimensions of a test fixture you presently have. The objective is to make measurements of a cavity-backed slot which should eliminate the problems you have experienced earlier with the excitation of parallel plate waveguide modes in the stripline feed of the fixure. This modification also has the advantage of more closely matching fixture measurements with the numerical modelling capability so that verification can be made with greater confidence. This memo describes dimensions and geometry of the proposed additions to the fixture based on a brief numerical study. As I recall, your fixture allows the radiating slot to couple directly to free space or to an intervening section of rectangular waveguide. Since the direct case eliminates some of the numerical code which remains to be verified, it would be appropriate to begin the measurements with this case. Once we can establish the validity of the model for this case, we can proceed to the indirect case and/or covered slots. Two dimensions were discussed at the meeting; the slot offset to obtain a matched element, and the proximity of the side walls to the strip at the cavity feed-thru aperture. The key considerations relevant to the latter are illustrated in Figure 1. In order not to disturb the dominant stripline mode, both as it enters and exits the cavity, the wall edges must not significantly disturb this field distribution. At the same time, for the cavity to have a predictable influence on the slot, the walls must appear as effective short circuits to the higher order modes, especially the 1t order mode which is much like the TEo1 waveguide mode. We therefore want to keep the aperture as narrow as possible but wide enough avoid interaction with the dominant mode. As can be seen, reducing the ground plane spacing provides some advantage. From this figure, we conclude that if we use the.0625 inch ground plane spacing, our aperture should extend from approximately.100 to.280 inches ( somewhat arbitrarily taking 10 % of the maximum field value as the limit and allowing for some manufacturing and assembly tolerance). For this ground plane separation, a strip width of.050 inches gives a line impedance of approximately 50Q, the dimension used for the data in the figure. In order for the resonant slot length to be less than the width of the cavity ( the dimension transverse to the line ), the width is dimensioned to allow the 1st higher order mode to propagate. The cavity length then has a strong influence on the resonant length and can be used to set it to a desirable value. The slot offset also

Comparison of Stripline Fields vs. Ground Plane Spacing 1.00.,,,,,........... 0.90 0.80 0.70 C 0 Spacing =.125 inches 0.60. 0 5 iSpacing =.0625 inches 0.50 o 0.40 0.30 0.20 E 0.10 Z oI. Io 0.000 0.100 0.200 0.300 0.400 0.500 0.600 0.700 Shielded Stripline Transverse Coordinate (inches) Figure 1: Dominant mode stripline field for two different ground plane spacings. has some affect on the resonant length so that these these two parameters together determine the equivalent slot impedance which represents a series transmission line element. Table 1 summarizes a set of dimensions which provides the desired slot impedance with reasonable geometry Figure 2 shows some ideas about a possible approach to adding the additional components needed to the existing fixture. The slot plate would be the existing pieces you may already have or the slot could be etched in the top plane of the stripline assembly. A change in slot length will move the resonant frequency which is acceptable since the program can be rerun at the new frequency point for verification. If the slot plates are used, the top of the cavity could be etched off the stripline assembly and the offset could then be varied by moving the stripline assembly relative to the slot plate in the fixture. You would probably need some kind of indexing scheme to get accurate offset settings. The stripline could be made the usual way except with notches cut out to accept the cavity walls. Another approach would be to use closely spaced'Z-wires' to form the cavity walls but this might be more tedious. These are all just casual ideas since I'm sure you have probably already thought about how to do it and undoubtedly have more experience in building these kinds of fixtures. Figures 3 and 4 show the affect of the offset on the coupling and resonant

Electrical Parameters Frequency 9.7 GHz Dielectric |r -'2.2 MIechanical Parameters (inches) Ground Plane Spacing (c) 0.0625 Strip WVidth (W) 0.050 Cavity WVidth (b) 0.750 Cavity Length (a) 0.650 Strip Centerline (yO) 0.1875 Slot Length (Ls) 0.4225 Slot W-Vidth (WVs) 0.050 Slot Offset (s) 0.0455 (Strip Centerline - Slot End) Table 1: Parameters for test fixture cavity-backed slot. length, respectively. It is these curves we should verify initially before proceeding to the waveguide coupled case ( the.0625 inch spacing case is the preferred case but not essential ).

Hold -down I Indexing bar Cavity Cap Slt iy Slot Platt Slot critr f ett Fi igure'l:'Saiil)le apprl,oacl for fixture modification.

CoLupling: Normalized Resistance vs. Offset 1.80 - i.n _,, / 1.60 7 {_) 0.8( 7 - 0.60 -, ().40 0.20 ()~00.... I,,........ I... I.... 0.000 0.010 0.020 0.030 0.040 0.050 0.060 0.070 0.080 SloL Offsct (inches) Spacing =.125 inches, Ws =.10 inches....... Spacing =.0625 inches, Ws.10 inches.... — Spacing =.0625 inches, Ws -.05 inches Figure 3: Slot coupllillg: iNor'malized I)eak resistance vs. slot offset.

Zero Reactance Length vs. Offset 0.45. I, I,. I. 0.44 ().44 0.44 0.43.. 0.42 0.410.41 - ". - 0.41 ().4().. O.((X) 0.010 0.020 0.030 0.040 0.050 0.060 0.070 0.080 Slot Offset (inches) Spacing =.125 inches, Ws =.10 inches. Spacing =.0625 inches, Ws =.10 inches ~....- Spacing *.0625 inches, Ws =.05 inches Figure 4: Resonant length vs. offset. 6

APPENDIX G Norman L.VandenBerg Investigation of a Stripline Slot Coupler January 16, 1991

An Investigation of a Stripline Slot Coupler N. L. VandenBerg and P. B. Katehi Radiation Laboratory, The University of Michigan, Ann Arbor, MI (January 16, 1991) Background Since our last contact late in December, we have generated some data based on our model of aperture-coupled microstrip lines using the dimensions you have specified. For reference, the geometry is duplicated in Figure 1.400" ~180".200.1o0".0472"..0472 ScmpLin e Codndc r s a # 1 Coupling Apamo in Figure\ Common G trd Pls o 2 ~\ ~(.200'x.407 / Aswehavdiscusse previouslyourmodelisbaed anOassumption of narrow031'.031s c o.O'd _." i.03i..062".03r1. E r 233 Coppa Tlicbms, =.0007 Figure 1: Geometry for stripline-fed slot coupler. As we have discussed previously, our model is based on an assumption of narrow slots. This is a convenience which requires us to model only one component of the field in the slot, simplifying the numerical implementation. Since the aspect ratio for your geometry is rather low, it is not obvious whether this assumption can be made or not, therefore we must allow the potential for some inaccuracy in the results. On the other hand, the problem has a great deal of symmetry

and arguments can be made for neglecting the transverse electric field component which is omitted in the present model, so we can still be somewhat optimistic. \We emphasize that this limitation applies only to the current implementation and not to the general formulation. If necessary, the other component could be added if a compelling need arises, however, this would take some amount of effort which is not currently planned. A fact that we also discussed is that our model is based on a shielded structure. With the current implementation, the dimensions for the shield are required to be such that the higher order modes of the stripline are cut-off. Again, this limitation is a consequence of convenience and is not a true limitation in the sense that a more flexible program is possible, but not currently available. Of course, this is not a liritation in many cases since we would often expect to analyse these types of couplers in some kind of packaging where we would not want higher-order modes propagating anyway. Indeed, this was the original intent for this program. where now we are attempting to use it for a slightly different purpose. We have also implemented a more general case for radiating slots without this kind of restriction but again, a more general implementation of the coupler program is not currently planned. The potential affect on the data to be included here are as follows: 1. A given slot length - that is, the dimension orthogonal to the axis of the strip - puts a minimum bound on the width of the shielding structure. (We assume the ground plane spacing is smaller than the width to the side walls and thus is not the limiting factor). This produces an upper bound in frequency since this dimension is directly related to the cut-off frequency for the first higherorder mode: for stripline, on the order of one-half the material wavelength. 2. To maximize high frequency capability of the model, the shield is usually chosen to be rather close to the ends of the slot if an actual dimension is not specified. This can have some affect on the response of the coupler due to the fringing effects of the fields at the ends of the slot. Therefore some caution is advisable in making a comparison of the results of this study to measurements on a structure which is not quite the same. Nevertheless, we would expect good agreement. Finally, upon seeing your drawing, we immediately noticed the steps in line width which you explained were intended to be part of an quarter-wave impedance transformer scheme. Now, although full-wave modelling of these types of steps is quite straightforward and in fact, we have previously done it in different contexts, this capability is not built into the coupler programs. Consequently, the data we provide will not include the affect of the step discontinuities. However, since the

steps are not very close, electrically, to the slot transition, it may be reasonable to model these transformers using conventional transmission line analysis. At the given frequencies, available CAD programs should provide acceptable performance for these types of steps so it should be relatively simple to produce a set of data comprised of the results we provide for the slot coupling and CAD models for the step transformers. For this purpose, we will provide the numerical data corresponding to the results shown here if desired. The most expedient method for transferring the data is by'E-mail' or'remote ftp' but it can also be supplied on diskettes if necessary. Results Before discussing the results, let us first define some terms and conventions: Phase Reference: In all cases we take the phase reference as center of the slot. (Not needed for discussion here but need to interpret the actual data to be supplied later). Line Stub Length (L): This length will be the length to which the stripline extends beyond the center of the slot which is the phase reference point. Slot Length (L.): The length of the slot is measured orthogonal to the longitudinal axes of the striplines. Slot Width (W.): The length of the slot in the direction of the stripline axes. As stated, to analyze the coupler shown we are ignoring the step transitions. We therefore will initially use a strip width of.180 inches. This corresponds to a 19.5~ line at 10 GHz. Because of the techniques used, in order to generate a frequency response plot, the program is run at each frequency of interest. For each frequency, it is relatively easy to generate a database which has the lengths of the slot and line stubs as parameters. For example, a typical output of a set of runs for 10 GHz is shown in Figure 2. A collection of data for the other frequencies of interest produces similar results. We then use a seperate program to scan this database to assemble a frequency response for a given set of line stub and slot lengths. The results for the present case is shown in Figure 3. As can be seen, the solid line, labelled L =.100 in., predicts a peak in the frequency response at about 11

Coupling vs. Line Stub Length vs. Slot Length 0.000 %)~ ~ ~ ~ 4 L, -.192 in..... L =.204 in. 3.4 FA ~-0 0L, O.216 in. Iinc Lcngth L a.228 in. I:~-J ~ ~ ~. L -.240 in.:3 -10.0 L -..25 2in. AL, a.264in. Lo -.276 in.;il I I ---- L,.Z2 in. L, -.300 in. -20.0. -.100 0.000 0.100 0.200 0.300 0.400 0.500 0.600 Line Stub Length (inches) Figure 2: Coupling versus line stub length and slot length for the 19.512 lines at 10 GHz. S21 Frequency Response -3.0 -6.0 -9.0 L =.100 in. -—.m. 1 ~ L =.191 in. -12.0 —.,- L =.260 in. -15.0 8.0 9.0 10.0 11.0 12.0 Frequency (GHz) Figure 3: Frequency responses for the 19.5fl coupler.

Coupling vs. Line Stub Length vs. Slot Length 0.000 - -—.. —.-. —- 1....- —... —-- L, I.192 in. -20.0 =-....204 n. ----- L, =.216zn....... L, -.228in. -410.0 0.000 0.I 0.200 0.300 0.600 L,.240 in Ls -.264 in. Ls -.276 in. Lo-..2 i. Ls -=.300in. -20.0 -.100 0.000 0.100 0.200 0.300 0.400 0.500 0.600 Line Stub Length (inches) Figure 4: Coupling versus line stub length and slot length for the 49.5Q lines at 10 GHz. GHz with about a 3 dB loss at 10 GHz. Note that a line stub length (L) of.100 inch corresponds to the dimensions given in Figure 1 which evidently has already been built and measured. We are interested to know how this compares to the measured results - don't forget though that the step transformers have not been accounted for in this result. The other two curves in Figure 3 show the influence of the line stub length on the frequency response. Based on this behavior, one would conclude that the stubs in the structure are too short for an optimum response at 10 GHz. We estimate that the proper length would be approximately.200 inches, assuming the step transformers do not significantly alter the behavior. To avoid the uncertainties introduced by not modelling the transformers, we have also analyzed the identical structure, except the line widths have been changed to.050 inches corresponding to 49.5Q line impedances at 10 GHz. A typical response versus line stub length and slot length is shown in Figure 4, again at 10 GHz. The rest of the database was assembled in the same manner and used to construct the frequency response plot shown in Figure 5. You may have noticed from the figures, that the slot length can also play a role in changing the frequency response. This is also true of the slot width and other parameters of the structure. A similar database can be formed based on these parameters and a similar study could be assembled. Hopefully, the results we have

S21 Frequency Response 3.0......... -3.0 -6.0 L L=.191 in. -9.0 --4a L =.214 in. — a,. — L =.237 in. -12.0 IWI L =.260in. -15.0 8.0 9.0 10.0 11.0 12.0 Frequency (GHz) Figure 5: Frequency responses for the 49.5f1 coupler. shown here will be helpful in your project. We also are interested in seeing how these results compare with the measurements. If there are discrepencies, we may be able and are willing to assist you in determining the causes.

APPENDIX H Norman L.VandenBerg Report of Progress Analysis of Finite Thickness Slots January 29, 1991

Report of Progress on Analysis of Finite Thickness Slots This note is to inform you of some preliminary work which has been done on the evaluation of the stripline-to-radiating-slot coupling problem for finite thickness slots. This geometry was illustrated in the original proposal as Figure la and is repeated here as Figure 1. To develop and test the initial modifications to the existing codes. we have simplified the problem to the case illustrated in Figure 2. Several additional subroutines have been added to account for the finite slot thickness for this particular case. Radiating Slot Rectangular Coupling / Waveguide Slot Stripline Figure 1: Stripline excitation of a waveguide-fed slot. Thick Slot Stripline / Figure 2: Simplified version of Figure 1. As a preliminary test, we have analysed a geometry which is sinmilar to another case for which we have avialable experimental results. These results were available in the literature and were used for comparison to a simple model which was recently developed and described to you earlier. You have already received a preview copy of the paper describing this model which has now been published in [1]. The

experimental data and the codes used in [1] are based on a longitudinal slot in empty waveguide. Because of the differences in the programs involved, the case we have analysed for comparison here is different in that the slot is backed by a cavity instead of waveguide and is excited by a stripline instead of the dominant waveguide mode used in [1]. However, we would expect the general behavior of the two to be similar if the overall structures are made similar. To that end. swe have analysed the geometry shown in Figure 3 which has similar dimensions to the waveguide case of [1]. The results are tabulated in Table 1. Thick Slot Stripline Figure 3: Preliminary test case geometry for comparison to results in [1]. As can be seen from the table, the correction factor obtained is nearly the same as for the simple model. The close agreement suggest that the results are probably correct since we expect the general affect of having finite slot thickness to be similar for both cases and insensitive to the type of the excitation. At the same time we expect there to be slight differences due to the cavity walls and the presence of the strip. So far then, we can conclude that the feasibility of modelling these types of structures by the method we have proposed looks quite promising. We thus are ready to begin the analysis and comparison of your experimental data as soon as it is available, to further develop and substantiate these models. In the meantime, work is continuing on modifying the remaining programs to evaluate the other structures we have discussed. References [1] Katehi, P. B., "Dielectric-Covered Waveguide Longitudinal Slots with Finite Wall Thickness." IEEE Transactions on Antennas and Propagation Vol. AP-38. No. 7, July 1990, pp. 1039-1045. 2

Case [1] I Full Wave Frequency 4.8 GHz Dielectric Er =2.69-2; thickness = 0.062 inches Cover Slot Slot hickn0.06 inches Thickness Transverse iTransnverse 1.87 x.87 inches Dimensions Longitudinal Longiotudnal 00 4.00 inches Dimension Offset from 2 mm 0.0 inches Centerline L | 0.374 w/o correction 0.363 thin slot,~o 0.397 w/ correction 0.390 thick slot Correction rFarctior l6.15 % 7.44 % Factor Table 1: Comparison of simple model and proposed full-wave model.

APPENDIX I Norman L. VandenBerg Corrected Results for Stripline Slot Coupler March 7, 1991

Corrected Results for Stripline Slot Coupler N. L. VandenBerg and P. B. Katehi Radiation Laboratory, The University of Michigan, Ann Arbor, II (March 7. 1991) This paper reflects changes in the numerical results discussed in our previous communication dated January 11, 1991. As you already know, the changes are a results of a sign error in the program which we identified in the course of making our own measurements on a similar structure. Because of the good agreement we found in that case, we now feel that the present data should give a good indication of how the structure we have been discussing should perform. Of course, this statement must be qualified by the conditions which we have stated previously and which still apply, that is, the present geometry may not meet some of the conditions we have assumed, allowing us to neglect certain factors. Therefore, the potential for some discrepancies in the results for this case still exists and we would suspect these factors play a role in any inaccuracies that may be found. In general, the new results show that the current geometry does not seem capable of obtaining optimum coupling through the slot, i.e. S21 = O dB, at least within the variation of parameters used here. In this case, as before, we have varied two fundamental parameters; the slot length (L,) and line stub length (1). The effect of varying these parameters is demonstrated in Figure 1 which is similar to Figure 2 of our previous memo except for the peak level of coupling. You can see that the coupling as a function of line stub length behaves essentially the same as in our previous results. As the slot length (L.) is varied however, the peak obtainable coupling is increased for each successive lengthening of the slot, up to a peak coupling of about -1.5 dB at a slot length of.300 inches. Careful study of the results also shows that the curves are reaching the point of diminishing returns, that is, it does not seem that a further lengthening of the slot will ultimately lead to 0 dB coupling. In fact, most likely at some point the trend will reverse and coupling will be reduced based on our previous interpretations of the influence of slot length. As before, an ensemble of similar curves is generated at different frequencies and a frequency response is obtained. For the 19.5Q lines, the result is shown in Figure 2 for several line stub lengths with the slot length set to.240 inches corresponding to the dimension used in the experiments. Again the curve for I =.100 inches reflects the case which was originally built. The I =.200 inches case is the dimension we suggested in our previous report.

Equivalent Circuit and Expected Result of Measurements The equivalent circuit of Figure 1 is also applicable to the T-Bar or open-end cases. The difference is that the load, ZL, is actually inside the cavity and the cavity itself then looks like a one-port element. Because the cross-bar is shorted to the walls, ZL presents a purely reactive load to the slot. (This is only true to the extent that we can neglect the coupling between the cross-bar and the slot which is what we had originally planned to do. It is probably a good approximation to first order however, as I mentioned in our conversation, we have expanded the model to include this coupling to remove any uncertainty about this factor). Thus the one-port impedance of the slot, referenced to the slot location, is simply the sum of the slot impedance and the load impedance referenced to the slot plane. De-embedding the slot impedance from the total measurement then is reduced to accurately determining the reference plane and.oad impedance. If the reference plane can be accurately set, the load impedance is easy to measure by closing the slot with a conductor and making the measurement. (Copper tape can be used or an identical test piece can be built without the slot). An alternative is to simply measure the element as you have done and consider the total impedance as the slot impedance, without regard to the separate components inside. This is equally acceptable and at the moment we don't know whether either approach is to be preferred over the other, although this latter approach is conceptually simpler. Perhaps you have a preference or convention. Neither eliminates the difficulties discussed above since the fundamental quantity we derive is r. Plan of Action The above discussion gives some background which outlines the approach for analysis of the two-port slots. Also included is some discussion of the difficulties encountered in determining the impedance of the T-Bar fed slots. To resolve these problems, we have not yet looked at the current formulation long enough. to know whether the current approach can work or not. We also have in mind, several alternative approaches which can potentially lead to a solution, each requiring a different amount of effort and each, as always, having their own uncertainties as well. In the coming weeks we will continue to look at the present formulation to determine whether or not the correct impedance values can be derived. If not, we will then re-evaluate the situation and explore the other alternatives.

10 GHz Coupling vs. Line Stub Length "s. Slot Length 0.0,....,... L,=.207 in. o.. L, L.217 in. L,s M.228 in....... L, =.238 in...L, =.248 in.' -10.0 L... L.259'M.. LI =.269 in. L, -.279 in. L, =.290 in. L,a.300 in. -20.0 -0.100 0.000 0.100 0.200 0.300 0.400 0.500 0.600 Line Stub Length (inches) Figure 1: Coupling versus line stub length and slot length for the 19.5fl lines at i0 GHz. S2i Frequency Response for 19.5f Lines vs. Line Stub Length -10.0 / A,, / / / -20.0 - - -l - - 40 50 6.0 7.0 8.0 9.0 1O 11.0 12.0 Finquncwy (GHz) Figure 2: Frequency responses for the 19.51' coupler.

S21 Frequency Response for 49.5Q Lines vs. Line Stub Length 0.0 - -—....- -?'"- - - -'-r ~... 1...... - - 1'' -'' -20.0 -'''/'.t...... 4.0 5 6 7.0 0 9.0 10.0 1.0 12.0 r-us (GHz) Figure 3: Frequency responses for the 49.5Q2 coupler. A similar procedure was used to generate a frequency response for the 50Q lines which is shown in Figure 3. You'll notice that the curves have substantially the same shape for this case except the peak coupling is about 1.5 dB lower. Obviously, narrower line widths do not improve the peak coupling but degrade it. Remember that for both cases we have again used uniform lines with no step transformers as explained before. One final note on the data which is a reminder of the discussion in the previous report. All cases here were generated using a box width of.300 inches, corresponding to a cut-off frequency of approximately 12.9 GHz for the higher order modes. Consequently, since the method of extracting the circuit parameters assumes only one mode is present, the data becomes more and more suspect as this frequency is approached. Therefore, some judgement is required when considering the agreement with experimental data at the high frequency end. In conclusion, we have seen some changes in the results generated in the previous report due to the discovery of a sign error in the program. This has changed the shape of the expected frequency responses somewhat, the most significant change being the introduction of a 2 to 4 dB loss in the peak coupling. The general behavior however, seems to be substantially the same as what we have seen previously, except for the loss. Due to the limited gains achieved by lengthening the slot, we suspect that the slot width may now be the limiting factor, although a wider line may improve peak

coupling as well. In fact, by narrowing the slot width, this case becomes closer to the one we have measured for which we obtained O0 dB coupling over a broad frequency range. Another remaining possibility is that the assumptions we have made concerning the aspect ratio of the slot dimensions may be invalid as discussed in the earlier report. As you may recall, we explained that our current programs are based on an assumption of narrow slot widths for which the transverse component of equivalent magnetic current in the slot may be neglected. Since the ratio of length to width for the proposed slot may not meet this criterion, the assumption may be invalid in this case which may account for any remaining discrepancy between the results shown here and the experimental measurements. We hope that these curves now indicate better agreement with the experimental data.

UNIVERSITY OF MICHIGAN 11111111111111111111111111115111111 39:j 02384 no2662R