THE UNIVERSITY OF MICHIGAN 7692-3-Q Electromagnetic Coupling Reduction Techniques Third Quarterly Report 15 June 1966 - 14 August 1966 By J.A.M. Lyon, N.G. Alexopoulos, A.G. Cha C.J. Digenis, and W.W. Parker August 1966 Contract No. AF 33(615)-3371 Project 4357, Task 435709 This document is subject to special export controls and each transmittal to foreign governments or foreign nationals may be made only with prior approval of AFAL(AVPT), Wright-Patterson AFB, Ohio. Prepared for Air Force Avionics Laboratory United States Air Force, AFSC Wright-Patterson AFB, Ohio 45433

THE UNIVERSITY OF MICHIGAN 7692-3-Q FOREWORD This report 7692-3-Q was prepared by The University of Michigan, Ann Arbor, Michigan, under the direction of Professor Ralph E. Hiatt and Professor John A. M. Lyon and on Air Force Contract AF 33(615)-3371 under Task No. 435709 of Project 4357 (U) "Electromagnetic Coupling Reduction Techniques". The work was administered under the direction of the Air Force Avionics Laboratory, Electronic Warfare Division, Research and Technology Division, Wright-Patterson Air Force Base Ohio. The Task Engineer was Mr. Olin E. Horton, the Project Engineer Mr. Herbert Bartman. This report covers the period 15 June 1966 through 14 August 1966.

THE UNIVERSITY OF MICHIGAN 7692-3-Q TABLE OF CONTENTS Page LIST OF FIGURES iv ABSTRACT vi I INTRODUCTION 1 II EXPERIMENTAL STUDIES 3 2.1 Introductory Remarks 3 2.2 Decoupling of Two Antennas by Means of Absorbing Materials 4 2.2.1 E-Sectoral Horn 4 2.2.2 Slots 9 2.3 Decoupling of Two Antennas by Means of Chokes and Corrugations 12 2.3.1 Chokes 12 2.3.2 Corrugations 16 2.4 Decoupling of Two Antennas by Means of Parasitic Elements 18 2, 5 Decoupling of Two Antennas by Means of an RF Bridge 21 2. 5. 1 A Microwave Bridge to Decouple Two Slot Antennas (XBand) 27 2.5. 2 Proposed Improvements of the X-Band Microwave Bridge 32 2.6 Coupling Between a Spiral and a Slot 40 III ABSORBING MATERIALS 47 IV CONCLUSIONS 54 V FUTURE EFFORT 57 ACKNOWLEDGEMENTS 57 REFERENCES 58 iii

THE UNIVERSITY OF MICHIGAN 7692-3-Q LIST OF FIGURES Figure Page 2-1 Block Diagram of Swept Frequency Circuit 5 2-2 Recessed E-Sectoral Horn 6 2-3 Maximum Gain Versus Frequency for E-Sectoral Horns 7 2-4 Radiation Patterns of E-Sectoral Horns 7-8 2-5 Coupling' Versus Frequency for E-Sectoral Horns 10 2-6 Slot in Cavity with Absorber 11 2-7 Coupling Versus Frequency for Slots with Absorber 13 2-8 Maximum Gain Versus Frequency for Slots with Absorber 13 2-9 Radiation Patterns for Slots with Absorber 14 2-10 Coupling Versus Frequency for Slots with Absorber 14 2-11 Maximum Gain Versus Frequency for Slots with Chokes 15 2-12 Radiation Patterns for Slots with Chokes 17 2-13 Coupling Versus Frequency for Slots with Chokes 17 2-14 Radiation Patterns for Slots with Corrugations 19 2-15 Coupling Versus Frequency for Slots with Corrugations 20 2-16 Radiation Patterns for Slot with One Parasitic Slot 22-23 2-17 Radiation Pattern for Slot with Two Parasitic Slots 23 2-18 Basic RF Bridge 25 2-19 Wave at Receiver 25 2-20 Effect of Path Length Difference on Decoupling 28 2-21 Microwave Bridge 29 2-22 Decoupling by Bridge 33 2-23 Decoupling by Bridge 34 2-24 Decoupling by Bridge 35 iv

THE UNIVERSITY OF MICHIGAN 7692-3-Q LIST OF FIGURES (continued) Figure Page 2-25 Decoupling by Bridge 36 2-26 Decoupling by Bridge 37 2-27 Broadbanding Bridge by Two Links with Filters 39 2-28 Geometry of Slot and Spiral Antennas 41 2-29 Coupling Between a Circular Spiral and a Slot 42 2-30 Coupling Between a Circular Spiral and a Slot 43 2-31 Coupling Between a Circular Spiral and a Slot 44 2-32 Coupling Between a Circular Spiral and a Slot 45 3-1 Magnetic Q Versus Frequency 48 3-2 Loss Tangent Versus Frequency 49 3-3 Magnetic Q Versus Frequency 50 3-4 Magnetic Loss Tangent Versus Frequency 51 3-5 Magnetic Q and Loss Tangent Versus Frequency 52

THE UNIVERSITY OF MICHIGAN 7692-3-Q ABSTRACT During this report period, additional studies were made on horns and slots surrounded by absorbing material. New designs were used which permitted all of the absorbing material to be flush with the conducting surface. Swept frequency methods were used in obtaining values for the increased isolation over a considerable range of frequency. Studies were made on the isolation of two antennas through the use of chokes. A slot antenna was constructed surrounded by four chokes in the form of circumferential trenches. For two systems, each equipped with such circumferential chokes, an additional 22 db of isolation was obtained. Additional studies were made on the use of corrugations to increase isolation. The corrugations described here are not of an optimized design and do not use absorber material as has been planned for the future. Simple parasitic antenna elements, such as slots as reflectors, have been used to improve system isolation. The improvement is rather small, being limited to about 7 db. Considerable effort has been utilized on developing an RF bridge to improve isolation by cancellation of the unwanted signal. Results have been encouraging but work remains to adapt this method to a wider bandwidth.

THE UNIVERSITY OF MICHIGAN 7692-3-Q INTRODUCTION Several of the items covered during this report are continuations of studies commenced and partially described in the previous quarterly report. Emphasis has been placed upon obtaining as much bandwidth as possible for decoupling for each of the decoupling methods studied. For convenience the measurements were often made at the X-band frequencies but none of the decoupling methods is limited to that frequency band. The initial experimentation using absorber was sufficiently encouraging so that newly designed sectoral horns and slot antennas were used in connection with absorber. The absorber studied was Eccosorb MF-124. Systems, each of which used an E-sectoral horn, were found to have an increased isolation of 18 db. Two systems each using a slot antenna, with surrounding absorber as studied here, show an increase of 13 db isolation. In both instances, the increased isolation can be obtained over the entire X band. A slot antenna was designed with four circumferential slots surrounding the driven slot. Two systems, each having such a connected slot, with surrounding circumferential chokes or trenches, would have increased isolation from one system to another of the order of 22 db over 0. 5 GHz within the X-band of frequencies. Two ordinary slot antennas were tested for increased isolation from one to another by the insertion of corrugations between the slots. The tops of the corrugations were flush mounted with the conducting surface. The systems were then found to be isolated by an increase of 5 db over the entire X-band. The work on corrugations is by no means complete; it is expected that corrugations of different depths

THE UNIVERSITY OF MICHIGAN 7692-3-Q and with different loading materials in the trenches will be used. Some favorable results were obtained through the use of parasitic antennas. In the experimental case explored, a slot antenna was considered with a second parasitic slot in back of it. The parasitic slot acted as a reflector. If the transmission systemkhas a slot radiator with a slot reflector and likewise the receiving system has a slot antenna with a slot reflector, it is found that the isolation increase between the two systems using these parasitics is approximately 7 db. Additional parasitic elements used as directors had very little effect as far as increase of isolation. The isolation increase was good for a 2. 5 GHz bandwidth for X-band operation. Another method of isolation explored, at least in a preliminary way, was throng the use of a bridge link or cancellation method. The principle of the bridge link is to make a second path from transmitting antenna to receiving antenna which will deposit a signal at the receiving antenna which will just cancel the undesired signal from the transmitting antenna. Proceeding in this fashion, a bridge method is described in the report, wherein 15 db of increased isolation is obtained for a bandwidth of 1. 5 GHz for X-band operation. This seemed to be very promising. Extensions of the bridge method, including filtering, are described later in the report.

THE UNIVERSITY OF MICHIGAN 7692-3-Q II EXPERIMENTAL STUDIES 2. 1 Introductory Remarks In studying methods for achieving reduction of the electromagnetic coupling between two antennas mounted on a common ground plane the following information in terms of charts or graphs is considered to be necessary and sufficient in most cases: maximum gain versus frequency, E- and H-plane radiation patterns at one or more frequencies, E- and H-plane coupling versus frequency, and E- and H-plane coupling versus receiving antenna orientation at one or more frequencies. It should be pointed out that the term "gain" is used with the meaning (Kraus, 1950) "directivity times efficiency". This clarification becomes necessary because some authors use the term gain with the meaning of directivity. When absorbing materials are used in the vicinity of the aperture of a microwave antenna with the purpose of reducing the interference to a nearby antenna it has been observed (Lyon, et al, 1966) that in some cases although the antenna directivity is not seriously affected, the antenna efficiency is reduced resulting in considerable loss of gain. Since any decoupling method resulting in gain reduction is not considered an acceptable solution the variation of gain must always be studied. The types of antennas considered radiate mainly broadside to the conducting ground plane. Therefore, the variation of gain was studied in this direction. Coupling from one antenna to another when the receiving antenna is rotated has already been presented (Lyon, et al, 1966) Since for aerodynamic reasons all antennas on aircraft or aerospace vehicles are flush-mounted any coupling reduction technique would not be acceptable unless if it avoids any protrusions above the ground plane. All techniques presented in this report meet this requirement.

THE UNIVERSITY OF MICHIGAN 7692-3-Q One important aspect of decoupling is broadbanding. A swept frequency technique was used to give faster and more complete information on the dependence of coupling upon frequency than the previously used point-by-point method. The circuit, in block diagram form, is shown on Fig. 2-1: When sweeping the frequency one has to abandon the slide-screw tuners used in the point-by-point method to match the aperture with the waveguide impedance. By using the tuning devices, the coupling obtained represents the most pessimistic estimate. However, the elimination of the tuners was not found to be responsible for more than approximately 0. 5 db change in the coupling level; besides the new system most probably represents a more realistic case in view of the applications. When measuring coupling, flanges A and B (Fig. 2-1) mate to establish the reference level shown on all swept-frequency coupling charts. In measuring variation of gain versus frequency a standard 2" x 2" horn was used as a receiver (inthe far-field of the antenna). The gain patterns indicate only the relative level, i.e. whether a particular decoupling method tends to increase or decrease the gain of the original antenna. 2. 2 Decoupling of Two Antennas by Means of Absorbing Materials 2.2.1 E-Sectoral Horns A new arrangement was designed and manufactured to permit the aperture of an E -Sectoral Horn to be surrounded by absorbing material and at the same time result in a flush-mounted structure (see Fig. 2-2). The absorber used is Emerson and Cuming's Eccosorb MF-124 for which the manufacturer specifies an average value of attenuation 69 db/cm (at 10 GHz). The variation of gain with frequency is shown in Fig. 2-3. The comparison with the control horn shows a small loss in antenna gain of the order of 0.5 db. Radiation patterns for three different frequencies are shown in Fig. 2-4. The pattern at 10 GHz, where the strongest coupling has been observed, shows that the

Sweep Oscillato Precision Oscillator Ferrite 8 - 12.4 GHz Attenuator Flexible Waveguide Isolator XMTR gu~~~~~~~~~~~~~~E HP x382 A Alfred 645KB HPx382 Crystal Ferrite Detector Isolator RCVR HP 423 A ~~Chart PositionGround rJI C hart ~~~~~~~~~~~~~~P os iti one r Cryst. -Bolom. Pen Function Recorder oto Plane ) Recorder ~~Control Amplifier Amplifier Sci. Atlanta Sci. Atlanta Sci. Afianta ScAtlnaPFA 23A APR 20 PC - 1 CBA 21 R-9Crystal 21 2"x 2" Detector RCVR Standard HP 423 A Horn FIG. 2-1: BLOCK DIAGRAM OF SWEPT FREQUENCY CIRCUIT.

=' ~X-GRO BAND PLANE 9.01~~,] FIG. 2-2: RECESSEDI) E-SECTORAL HORN AD ABSORBER GEOMETRY (Dimensions in cm) 2.3 X- BAND WAVEGUIDE FIG. 2-2' RECESSED E-SECTORAL HORN A~JD ABSORBER GEOMETRY (Dimensions in cm)

THE UNIVERSITY OF MICHIGAN 7692-3-Q,,, I, r!I HONS (-) C oControl Horn kl:~~ ~High Isolation Horn 8 9 10 11 12 Frequency GHz FIG. 2-3: MAXIMUM GAIN VERSUS FREQUENCY FOR E-SECTORAL HORNS il it t t. t~ It_ i-i- i t.._. t;\ i4 I } 1 1, I -20 IA 90~ 60~ 30~ 0~ 30~ 60~ 90~ 900 60~ 30~ 00 30~ 60~ 90~ Angle from Broadside 8. 23 GHz Angle from Broadside (a) E-Plane (b) H-Plane FIG. 2-4: E- AND H-PLANE RADIATION PATTERNS OF E-SECTORAL HORNS, ( —) Control Horn; (-.o) Isolation Horn.

THE UNIVERSITY OF MICHIGAN 7692-3-Q -of~~~~~~ ' I t t I- I I - 1-!, —. — _l I:,. ' -I:,.... - _ Fz-lo-410X~ F -.,.-~ -- --!- _.. 4- l__f. 0 0 m-20-2-e- < _ -20-g&- t ---1 1 \:2 2 - 0 1 -I I:: L — 1- -1- -30- - L 90~ 60~ 30~ 0~ 30~ 60~9 90 90 600 30~ 0~ 30~ 60~ 900 Angle from Broadside Angle from Broadside (c) E-Plane 10.03 GHz (d) H-Plane. + " i.;>-20 I' -3 o ~ ~ - )8 I @ ~' '~,: ~ ~ i:. IIi i i 411 9-0 60 30 0 3060 90 90 60 30 0 30.60 90 (e) E-Plane 12.03 GHz (fl H-Plane ", 6!t-44- 3 0- '<4 Ii 1; 90060030 0 30 60 90 90 60 30 00 300 60 900 Angle from Broadside Angle from Broadside (e) E-Plane 12.03 GHz (f) H-Plane FIG. 2-4: E- AND H-PLANE RADIATION PATTERNS OF E-SECTORAL HORNS; (- ) Control Horn; ( o ) Isolation Horn.

THE UNIVERSITY OF MICHIGAN 7692-3-Q presence of the absorber leaves the main lobe practically unchanged, while it reduces the side lobes by approximately 9 db. Coupling patterns versus frequency for both the modified (high isolation) horn and the control horn (i.e. a flush-mounted horn with aperture 2. 3 cm x 3. 9 cm) are shown in Fig. 2-5. It should be noted that the high-isolation modification was applied only to the transmitting horn, i.e., the same receiver was used for all the coupling patterns of Fig. 2-5. Therefore, according to the symmetry argument, twice as much decoupling should be expected if both transmitter and receiver were modified according to Fig. 2-2. Thus for E-plane coupling the increase in isolation is approximately 18 db while for H-plane, it is 15 db. Since the swept-frequency generator output varied with frequency (by + 0. 75 db) constant coupling levels are no longer horizontal. A reference line corresponding to -34 db below direct coupling is shown on top in Fig. 2-5. This reference level was obtained by connecting directly between flanges A and B (Fig. 2-1) while introducing 34 db attenuation by means of the precision attenuator. This attenuation was then removed from the system during the actual coupling measurement. Patterns of coupling versus receiver orientation were also taken but are not presented since they do not contain any new information beyond that previously published data (Lyon, Kalafus, et al, 1966), except of course for the overall coupling level. 2.2, 2 Slots A slot was recessed in a shallow cavity which was partially filled with absorber. The cavity dimensions and absorber geometry and type are the same as the ones used with the E-Sectoral Horn (Fig. 2-2). This modification was found to increase the maximum gain of the slot by approximately 0. 5 to 1.0 db throughout the X-band while decreasing the radiation along the surface of the ground plane. Typical radiation patterns taken at a frequency of 10 GHz are presented in Fig. 2-6.

THE UNIVERSITY OF MICHIGAN 7692-3-Q -34 db level -35 -40 'e -45 Control Horn E-Plane L) 5High Isolation Horn E-Plane Control Horn H-Plane -60 High Isolation Horn H-Plane 8 9 10 11 12 FIG. 2-5: E- AND H-PLANE COUPLING VERSUS FREQUENCY FOR TWO E-SECTORAL HORNS SPACED 11.43 cm. (Eccosorb MF-124 Absorber). 10

00 00 0o 300 30 300 00 b-db N N0 -2 db -5 M t z ~/~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~i 0~~~~~~~~~ -4~~~~~~~~~~~~~~9 -~ r,~~~~~~ Zd 60~~~~~~~~~~~~0~ ~0 -8 020 \1 cn GROUND ROUND '/ G(aPLANE PLANE b FABSORBER 2-6 9 ASO SOBER - N I- 0- o~ I I~ 3 -40 90 - 90~~~~~~~~~~~~~~~~~~~~~~~~~r C~ (a) E-Plane (b) H-Plane FIG. 2-6: SLOT IN CAVITY WITH ABSORBER (Eccosorb MF-124) ANTENNA GEOMETRY AND FAR-FIELD RADIATION PATTERNS AT 10 GHz (Dashed Lines Show Corresponding Radiation Patterns of Plain Slot Dimensions in cm).

THE UNIVERSITY OF MICHIGAN 7692-3-Q The E-plane coupling between a modified and a plain slot is shown in Fig. 2-7, where it is also compared to the coupling between two plain slots. A reduction of maximum coupling by 6. 5 db is observed over the X-band of frequencies. Again, by reciprocity, if both antennas were similarly modified, a 13 db decoupling would result. Next the absorber was changed to B. F. Goodrich's RF-X (same geometry). Variation of maximum gain and coupling with frequency as well as radiation patterns for this case are shown in Figs. 2-8 through 2-10. It should be noted that very similar patterns to those shown in Figs. 2-8 to 2-10 were observed when no absorber at all was placed in the cavity, indicating that the RF-X absorber is not suitable for this application. 2.3 Decoupling of Two Antennas by Means of Chokes and Corrugations 2.3.1 Choke s A new slot antenna was constructed having an X -band slot surrounded by four chokes in the form of circumferential trenches. If b is the radius of such a choke then the cutoff depth is given by X X d = g = o (X = free space wavelength) c 4 X 2o 2rb Since previous data (Lyon, et al, 1966) have indicated that for frequencies below the cutoff the gain of the antenna is decreased, while the coupling to nearby antennas is increased, the chokes were designed with a cutoff frequency of 8.2 GHz, The antenna gain was found to increase by 4 db at this frequency (Fig. 2-11) due to the deformation of the radiation pattern of the slot caused by the chokes (Fig. 2-12). The E -plane coupling between a slot surrounded by four chokes and a plain slot was 11. 5 db below the coupling between two plain slots at the cutoff frequency

THE UNIVERSITY OF MICHIGAN 7692-3-Q -20 20 db level ' Ii~;'ii i3il: i 1tX lili',:I i: r~ i " ' ilCoup i tBe twee.:i~L l-~-']r:'T i-,,._.,4_....| o:'i/i /tiit 1 il i i/ Two Plai n Slotsj ~_!.!- 1Coupling Between rou ~ ~ ~ ~ ~ ~ ~ ~ 4 _.Two Plain Slots -~o,-t-!,', —,,-l',,,,,,,,,,!' i-rtl-l-fi-r tH-:iii I~i!Coupling Betweoen A ModifieId and A "i!- iil i ~l il; II!- It/l/i ~111 i Ti F i 11 ii - IPlain Slot 8 9 10 11 12 Frequency GHz FIG. 2-7: E-PLANE COUPLING VERSUS FREQUENCY FOR TWO SLOTS SPACED 11.43 cm (Absorber Eccosorb MF-124). Generator Output Plain Slot Modified Slot / 8 9 10 11 12 Frequency GHz FIG. 2-8: MAXIMUM GAIN VERSUS FREQUENCY FOR SLOTS (Absorber: B.F. Goodrich RF-X). 139 -

THE UNIVERSITY OF MICHIGAN 7692-3-Q I 1!1 I I I 11 1 1 4 111 I!I -0 0i l ' ifi i 'L aPl)tLlti EPl a in i( H-P FIG 2:i E i AN HL RAIAO -- Pt S 10;,-10. I I I I I I N I I... o { i!! i i- 3.-3 Ai Ife d } J i ' { Modified a.. o Plain SlotF:11 llbikn Sot -1 —20 -20_- -20o 2 0 0 01 3 0 16 90~ 60~ 30~ 0~ 30 60~ 90 90~ 60~ 30~ 0~ 30~ 60~ 90~ (a) E-Plane (b) H-Plane FIG. 2-9: E- AND H-PLANE RADIATION PATTERNS AT 10 GHz. -25 / 14 -30 db level -30 PI I RICouping Between:@ | | \ | Two Plain Slots o 0 | '\1 / | Plain Slot -40 8.2 9 10 11 12 Frequency GHz FIG. '2-10: E-PLANE COUPLING VERSUS FREQUENCY FOR TWO SLOTS SPACED 11.43 cm. (Absorber: B. F. Goodrich RF-X). 14

o!=3~~~~~~ =-~~~~~~~ =Generator Output Form Ct3 Slot With Four Chokes Plain Slot 8 9 10 11 12 Frequency, GHz C FIG. 2-11: MAXIMUM GAIN VERSUS FREQUENCY FOR SLOTS.

THE UNIVERSITY OF MICHIGAN 7692-3-Q (Fig. 2-13). This decoupling, however, was gradually reduced at the higher frequencies. Thus the decoupling over the entire X-band, would be approximately 13 db if both slots were surrounded by chokes. The H-plane coupling was very little affected by the presence of chokes. The decoupling observed was of the order of 1 to 2 db. 2. 3. 2 Corrugations The propagation of an electromagnetic wave along an infinite corrugated surface has been studied (Hurd, 1954). The data in this reference show that the cutoff depth, d, depends upon the spacing of the corrugations, t. For a separation t< l then d =0. 23 X to 0. 25X. The surface wave does not propagate for d <d< 01 c o c 10' Three sets of corrugations have been used during this period. All three were imbedded in a cavity placed symmetrically between two slots, oriented for E-plane coupling. The aperture dimensions of this cavity were 7. 2 cm and 2. 3 cmparallel to the H- and E-plane respectively. The depth of the cavity was adjustable so that the corrugations would be always flush-mounted. The parameters of the sets of corrugations used are given below: Set I: d =0.9 cm t = 1. 5 mm Set II: d2=0.9cm t2 =2.1 mm Set III: d3 =0.6to0.9m t3 =1. 5 mm In the third set the depth was varied in equal steps monotonically, so that in an Eplane cross-section the bottom of the cavity would be in a ladder form. The sets I and II had almost identical radiation patterns (within 0. 2 db) which seemed to be greatly influenced by the existence of the cavity as shown in Fig. 2-14a, Set III created a radiation pattern of the same general shape but with 16

THE UNIVERSITY OF MICHIGAN 7692-3-Q T — 2 + e r + - ' tI. + _ _t I 1 it! 0//, — i-i $; P i n I I S I ( I I [ I St W ii Fu C- o k e II.,- ' '! I LiJ. -20 -20 ib level i 40,i —4 |.- Sloti 1 ~~- -10 11j+~ ii -1 I' I -{ -- - - -? i - - - -, -- > '., iit J. i - rtl~t,,_ Two Plain l o t 908 9 10 3011 12 FIG. 2-13: E-PLLA E RADIATION PATTERSUS ATFREQUENCY FOR TWO SLOTSGHz 17....

THE UNIVERSITY OF MICHIGAN 7692-3-Q different size side lobes (Fig. 2-14b). The maximum gain in the presence of the tapered corrugations was + 1 to -2 db with respect to the gain of the plain slot, depending upon the frequency. The decoupling obtained was of the order to 5 to 8 db (Fig. 2-15). In view of the decoupling levels observed and the accompanying reduction of gain at the broadside direction, this method at its present stage of development is not considered satisfactory for coupling reduction. It should be noted, however, that the tapered corrugations offer more decoupling without any additional loss in gain. The design of improved tapered corrugations with desirable a frequency characteristic will be attempted in the future. 2.4 Decoupling of Two Antennas by Means of Parasitic Elements Parasitic elements can be used as reflectors or directors to reflect or direct, respectively, the antenna radiation in a particular direction. Such elements have been found to be very effective at VHF and UHF frequencies (e. g., Yagi-Uda Array) and have numerous applications. The use of parasitic elements with microwave aerospace antennas is severely restricted by the requirement of flush-mounting. In initial experiments good results were obtained by using four screws errected on the ground plane betweentwo E-Sectoral Horns. In this case the E-plane coupling was reduced by 18 db over a frequency range of 0. 5 GHz (9. 25 to 9. 75 GHz). In order to study the behavior of flush-mounted parasitic antennas a new antenna has been constructed in the form of a three slot array. One slot was fed by X-band waveguide while the other two were backed by cavities constructed again from X-band waveguide and equipped with sliding shorts. The distance of the parasitic slots from the transmitting one was adjustable, from a minimum of 1. 27 cm center-to-center (imposed by the physical dimensions of the waveguide) to a maximum of approximately 4. 0 cm.

+5 +5 - ~ ~ ~ ~ ~- rr OESet III, ~r4~~~~~ - Deeper Near XMTR, j t f i | 9 S C C 0 R 0;Plain Slot 0 0 Pc;~ - t m 1 _ 5 [Set III, Shallower Near XMTR Corrugations, Set I I cD 1 0I 11 ' l |tt=, -1 4| I 10~~~~~~ t11 -lt t: IIt -I 90o 60 300 00 30~ 60~ 900~ 960 ~ 30 06 3 Angle from Broadside Angle from Broadside (a) (b) FIG. 2-14: E-PLANE RADIATION PATTERNS OF SLOT IN THE PRESENCE OF CAVITY AND CORRUGATIONS AT 10 GHz.

THE UNIVERSITY OF MICHIGAN 7692-3-Q -25 -30 db level 4 -30 - Corrugations, Set II -35. 8 9 10 — 1 12 Frequency GHz (a) -25 - Coupling Between Plain Slots -30 db level -30.F-4.~ | Coupling with Intervening Tapered Corrugations -35 Set III, -40 8.2 9 10 11 12 Frequency GHz (b) FIG. 2-15: COUPLING VERSUS FREQUENCY FOR TWO SLOTS SPACED 6.5 cm WITH AND WITHOUT INTERVENING CORRUGATIONS. 20

THE UNIVERSITY OF MICHIGAN 7692-3-Q The case does not lend itself readily for an exact theoretical analysis because the close spacing- of the parasitic elements necessitates solution of the combined problem, transmission line - - antenna, i.e., one would have to consider the field distribution in the aperture of the transmitting slot as unknown. If one were to use "thin" slots then the field distribution in the transmitting slot may be assumed known and on this basis the problem can be solved, with the aid of a digital computer (Coe, et al, 1964). A series of radiation patterns have been taken for various spacings, I, of the parasitic elements and various positions of the shorting plungers. A single parasitic element acts as a reflector when the cavity depth d is X /4. A front to back ratio of 7 db was observed for close spacings. For the minimum possible spacing the pattern becomes highly asymmetrical while for spacings near X /2 (center-tog center) the pattern symmetry is partially restored (Fig. 2-16). No setting was found for which a single parasitic element would act as "director", increasing the side lobe level in its direction. In combinations of two parasitic elements on either side of the transmitter it was noted that the "reflecting" parasitic had a predominant effect. A front-to-back ratio of 8 db has been obtained in this case (Fig. 2-17), Coupling measurements have not been completed yet. So far a modest decoupling of 3 db has been observed if one of the two slots of a transmitter-receiver system is accompanied by parasitic elements. Twice as much would be expected if both slots were similarly modified. It is felt that considerable improvement will be possible in the near future. 2. 5 Decoupling of Two Antennas by Means of an RF Bridge If two antennas are in proximity to each other, there may be strong interference or "coupling"between them. An RF bridge has been used to reduce this interference, or to "decouple" the two antennas. 21

(u1lovds aulnao-oj- a4uaoa = 7 ) 'ZHD OI ~LV LHDUI ItH, NO tiLOrIS oIIsvidv atNO IO 3oNaLS3acid aHIL NI LOrIS V JO SNIIaLLLVd NOIIVIVuI aINVIcd-a:9I-z 'DII U13 00' = P 'o t-l' = 7 (o) 006 009 0 o0 O0s 0 09 006 C 0 0 0- 0. L- t t- I,I F...-,,: i T,!.. '_I I. iF oo, p uio O I(q) I t - I- - - [ Ti-00 1=P 'v7 LZ1 7- 7 I ~ i T ~I I 006 009 oO0 00 o0s 09 006 IX~~~~~~~~g_ _ NVOIHOI 10 ALISIAI-i i, I -H

THE UNIVERSITY OF MICHIGAN 7692-3-Q 0- ----— t - _-__ 0 5 4- ce5 tr e r spa i-ng). MF-10F - 1F - I -' —1- --- -10... 90~ 60~ 30~ 0~ 30~ 60~ 900 (d) d = 2.00, d = 1.00 cm FIG. 2-16: E-PLANE RADIATION PATTERNS OF A SLOT IN THE PRESENCE OF ONE PARASITIC SLOT, ON THE RIGHT, AT 10 GHz. ( center-to-center spacing). -<< ~i I t l i -,i, l-+-~, l: I ~- i ~lil r -it-7. i t -! I t -I t I i 90 -1 60~ 3 0~ 30~ 60~ 90 Parasitic 1 Transmitter Parasitic 2 11 = o1 27 cm = 1.27 cm 2 d = 0.99 cm d2 = 1.98cm FIG. 2-17: E-PLANE RADIATION PATTERN OF A SLOT IN THE PRESENCE OF TWO PARASITIC SLOTS AT 10 GHz - 23

THE UNIVERSITY OF MICHIGAN 7692-3-Q The basic concept of the RF bridge is illustrated in Fig. 2-18. The interfering signal travels from the transmitter to the receiver along the upper path of length d1 cm (equivalent air path length). This signal path is called the coupled path. The RF bridge cancels out this interfering signal by producing a signal at the receiver which is of the same amplitude as the interfering signal, but 1800 out of phase with it, This canceling signal travels from the transmitter to the receiver along the lower path of length d2 cm (equivalent air path length). This signal path is called the bridge path and contains an adjustable phase shifter, labelled 0 in the figure. Consider first the coupled path. The number of waves which can be contained in a distance x is given by: N = x/X (2.1) where X is the wavelength. The wavelength function is determined by the medium through which the wave propagates. As a simplification distances d are expressed as equivalent air path lengths in cms. For free space: 10 = /f = 3x101(cm/sec) o f (Hertz) Now, if the transmitter is taken as the phase reference in Fig. 2-18, the wave arrives at the receiver along the coupled path with the phase: 01 (dl/o-nl) x 3600 (2.3) Where dl/X is the number of waves along the coupled path, and nl is the greatest integer < d1/X. The subtraction of n1 in Eq. (2. 3) is done so 00~ < 1 < 360. See Fig. 2-19 for a wave arriving at the receiver with phase 24

THE UNIVERSITY OF MICHIGAN 7692-3-Q (dlcm) Transmitter Receiver (d2 cm) 0 FIG. 2-18: BASIC R.F. BRIDGE Transmitter Receiver 1 180~ 00 )o3600 720 9Oo 0 o FIG. 2-19: WAVE ARRIVING AT RECEIVER WITH PHASE 0 = 1800 25

THE UNIVERSITY OF MICHIGAN 7692-3-Q 01 = 180 along the coupled path. In the figure, dl/X = 2.5 waves. Now consider the bridge path. It is similar to the coupled path except that it contains a phase-shifter which adds 0o to the phase angle in this path. (0 < 0 < 3600) Again, taking the transmitter as the phase reference, the wave arrives at the receiver along the bridge path with the phase: 02 (d2/X -n2) x 3600 + 0 (2.4) For the RF bridge to produce decoupling, the signal along the bridge path must cancel the signal from the coupled path. Thus, at the receiver: 01-02 = 180. (2.5) Combining Eqs. (2. 3) and (2. 4) yields: | (dl -d2) 360o_0o -n(3600) =1800 (2.6) where n is an integer. Obtaining broadband decoupling with the RF bridge is very important. The manner in which the decoupling varies with frequency must be known, Equation (2.6) provides much insight into this problem. In a physical example, the lengths of the coupled path and the bridge path will b fixed at d1 cm and at d2 cm respectively. The phase-shifter will be set at a phase angle of 00 to cancel the interfering signal at a frequency of fl Hertz, corresponding to a wavelength of X1 cm. Because Eq. (2. 6) deals with degrees, it is periodic with cycles of 360. Thus, the bridge will null at a wavelength of X., corresponding to f., when: 26

THE UNIVERSITY OF MICHIGAN 7692-3-Q l/Ax - l/Xi = K 1/(dl - d2) (2.7) where K takes on values of positive integers. Then, for a single setting of 0, the bridge will cancel the interfering signal at many frequencies. This happens whenever a frequency corresponds to an integral number of wavelengths, as shown in Eq. (2. 7). However, just as the bridge will null at many frequencies, there are also many frequencies where the bridge path's signal will add in-phase with the coupled signal at the receiver. This adding will actually increase the level of the interfering signal at those frequencies. This is undesirable. The difference in length between dl and d2 is the key to making the RF bridge feasible. If dl equals d2, the coefficient of the frequency dependent term in Eq. (2. 6) will be zero. For this case, 0 = 1800 is a requirement to null the bridge, independent of frequency. The decoupling obtained would be perfectly broadband. Moreover, there would be no in-phase adding of signals as described earlier. Figure 2-20 shows the relationship of dl to d2. When dl equals d2, the decoupling is broadband as discussed above. When the difference between dl and d is moderate, the nulls are fairly broad and moderately spaced in frequency. 2 When the difference between dl and d2 is large, the nulls are very narrow and closely spaced. Note the in-phase adding that actually increases the couplingwhich occurs whenever dl is not equal to d2. Thus, for the postulated RF bridge in Fig. 2-18, the decoupling obtained could be perfectly broadband if dl equals d2. 2.5.1 A Microwave Bridge to Decouple Two Slot Antennas (X-Band) The circuit of a microwave bridge used to decouple two slot antennas mounted on a common ground plane is shown in Fig. 2-21. This bridge circuit is designed

THE UNIVERSITY OF MICHIGAN 7692-3-Q Cou led Power Level Power 1 dl 2 Level in db Decoupled Power Level Frequency -- Coupled Power Level I dld2 | ft t \ / 0 / \Moderate Power Level _, Decoupled in db Power Level Frequency Coupled Power Level r I dd - d2 Large Power ~~~~~~~~Level ~Decoupled Power Level Frequency — FIG. 2-20: EFFECT OF Jdl - d2 ON DECOUPLING. 28

Slotted Line Alfred Swept 50 dbXbadGon Frequency Ioao Generator 1lotH 8-12 G~~~~~~~~~~~~z ~~~Transmitting Precision So Attenuator Flexible Waveguide (W) a2 Precision Attneuator Precision 143c Phase -Shifter 01.3c 3~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~ Hybrid Tee Matched Termination Isolator Slot B0 B~~~~~~ FIG. 2-21: MICROWAVE BRIDGE

THE UNIVERSITY OF MICHIGAN 7692-3-Q for X-band and has been built entirely with X-band waveguide components. RF power is supplied by an Alfred X-band swept-frequency generator. The input power level is adjusted by the precision attenuator ca1. It has been shown that broadband decoupling is possible with an RF bridge if the length of the coupled path is made equal to the length of the bridge path. The reference point which was the transmitter in Fig. 2-18 is the probe of the X-band slotted line in Fig. 2-21. The coupled path begins at the probe, includes the rest of the slotted line, the ferrite isolator A, the flexible waveguide section W1, the transmitting antenna slot A, the coupling path between slot A and slot B, the receiving antenna slot B, the ferrite isolator B, and the flexible waveguide section W2. The bridge path also begins at the probe of the slotted line. It includes the probe, the attenuator a2, the phase-shifter 0, and the section of flexible waveguide W3. The bridge signal and the coupled signal are added together in the hybrid tee. The resultant output passes through a crystal detector into the Scientific Atlanta receiver. The bridge path and the coupled path are made approximately equal in length by a careful choice of the lengths of the flexible waveguide sections. A fine adjustment of the coupled path length is provided by the slotted line. The probe rides in a movable carriage; the range of adjustment is 10 cm. The probe in the slotted line provides the signal for the bridge path. The precision attenuator a2 matches the level of the bridge signal to the level of the coupled signal. For the separation of the two slots shown, the loss of signal power is approximately 30 db between the transmitting slot and the receiving slot. The precision phase-shifter provides the necessary phase shift for the bridge signal to cancel the coupled signal at the hybrid tee. The ferrite isolators are used to prevent standing waves which might be caused by mismatches between the various components.

THE UNIVERSITY OF MICHIGAN 7692-3-Q An analysis of the microwave bridge circuit can be performed like the analysis of the general RF bridge. The expression for the arriving phase angle in the bridge path is basically the same as the earlier expression for 02 (Eq (2.4)). For waverl 2' guide, however, the relationship between the wavelength and the frequency is known: X = c/f x 1 - (c/f X )2 (2.8) g c where X is the cutoff wavelength = 4. 57 cm for X- band waveguide. Then, subc stitutingthis Xk into Eq. (2.4): 02 = (d2/c x fl1- (c/f ) n2) 3600 + 0 (2.9) where d2 is the length of the waveguide in the bridge path. In the coupled path, however, the wave travels through two different media. There is X-band waveguide, and there is the air path between slot A and slot B. Thus, there are two terms in the equation for the arriving phase angle: 01 = (dwg/X + dair/X -n1) x 3600 (2.10)1 where dw g is the length of waveguide in the coupled path and d air is the length of the air path between Slot A and slot B. Substituting the expressions for X (Eq. (2.8)) and X (Eq. (2. 2)) into Eq. (2.10) we get: 1 = ((dwg/c)x 1-(c/f+ (dair/c)x f-n1) 360 (2.11) 31

THE UNIVERSITY OF MICHIGAN 7692-3-Q Next, define a factor S, a function of frequency alone: S = f/c x - (c/f X )2 x 3600 (2. 12) As before, if the microwave bridge is to work, 0 and 02 must be 1800 apart. Expressed in an equation using the factor S: IS (dwg -d2 dair/ l - (c/f X )2) - n 360 = 180 (2.13) Equation (2.13) is similar to the RF bridge Eq. (2. 6). The factor S, which is a function of frequency only, is similar to (360 /X) in Eq. (2. 6). Before, we set dl equal to d2 and the coefficient of the frequency dependent term went to zero. We want to do the same thing in Eq. (2. 13). But looking above, we see that the coefficient of S has a part that is frequency dependent itself. Thus, the coefficient of S cannot be made to go to zero for all frequencies. Thus, we see that the microwave bridge of Fig. 2-21 does not produce perfectly broadband decoupling. However, the best performance would be obtained when the coefficient of S is as small as possible. Using the microwave bridge of Fig. 2-21, we have been able to obtain about 15 db of decoupling over a bandwidth of 1. 5 GHz in the X-band. The frequency where the decoupling occurs is variable in this band. Figures 2-22 through 2-25 illustrate the decoupling obtained with the microwave bridge in X-band. Figure 2-26 is included to show what happens if the coefficient of S in Eq. (2. 13) is too large. 2.5.2 Proposed Improvements of the X-Band Microwave Bridge In the precious section, the reason the decoupling was not more broadband is mainly the presence of the air coupling path between slot A and slot B. This 32

THE UNIVERSITY OF MICHIGAN 7692-3-Q Generator Power Coupled ' _5 | |Power -30 Decoupled -- Power -35 -40 8 9 10 11 12 Frequency GHz FIG. 2-22: DECOUPLING BY BRIDGE. 33

O, ~~i 1Coupled Power. -25' -30.: ~.> -35 li- Ti1.T a.).... I i I. 0 o T t iiii I __ 1 T45 J ~ - l t l 8 9 10 11 12 Frequency GHz FIG. 2-23: DECOUPLING BY BRIDGE, PI I co~~~~~~~~ -50 It~ cA3 ~ ~ ~ ~ ~ ~ ~ ~ ~ 1 11 1 3~~~~~~~~~~rqec G5 iz ~~~~~~I. 22:DCULIGB RDE

THE UNIVERSITY OF MICHIGAN 7692-3-Q 0 0 ~. ~~Generator Power -25 Coupled Power _300 Decoupled Power --- -35 --45 8 9 10 11 12 Frequency GHz FIG. 2-24: DECOUPLING BY BRIDGE 35

THE UNIVERSITY OF MICHIGAN 7692-3-Q 0 1 Generator - 25 Coupled -30 Decoupled -50 8 9 10 11 12 Frequency GHz FIG. 2-25: DECOUPLING BY BRIDGE 36

THE UNIVERSITY OF MICHIGAN 7692-3-Q -25 -30 MSACoupled Power -35 0d Qj-40 -45 Decoupled Power -50 8 9 10 11 12 Frequency GHz FIG. 2-26: DECOUPLING BY BRIDGE (PATHS MISMATCHED) 37

THE UNIVERSITY OF MICHIGAN 7692-3-Q air path produced a term in Eq. (2.13) which had a frequency dependent coefficient. One possible solution would be to place an equivalent air gap in the bridge path. This could be done with carefully aimed X-band horns, or with the proper length of airline coax. This would produce a term involving an air path in Eq. (2.13) which might be made to cancel the term present now. This change would give a chance of obtaining very broadband decoupling. The amount of decoupling obtained is dependent upon how well the amplitude of the bridge signal matches the amplitude of the coupled signal. If they were exactly equal, the decoupling expressed in db would be infinite. As the frequency is swept, however, the coupled power varies somewhat. Thus, to obtain more db's of decoupling, the bridge signal would have to match the variation of the coupled signal closely. We also need to investigate the properties of the components used in our system. We need to know how the phase-shifter performs as a function of frequency. The attenuator ca2 has some phase-shift inherent in it too. Another thing to investigate is the length of the coupling path between slot A and slot B as a function of frequency. In general, the response of our system components must be known before any small irregularities can be compensated away. A completely different idea is to achieve broadband decoupling by using two bridge paths and a filtering network. This is shown in Fig. 2-27. Bridge network 1 is set to null for the high frequencies; after passing through the high-pass filter the undesired adding in-phase at low frequencies is removed. Bridge network 2 is set to null at low frequencies; after passing through the low-pass filter the undesired adding in-phase at high frequencies is removed. The outputs of the two filters are added together, hopefully producing broadband decoupling. 38

THE UNIVERSITY OF MICHIGAN 7692-3-Q Coupled X ~~~~~Power ~Coupled Power~~~~Power High Pass Decoupled Filter Power'~ Decoupled Power Frequency —* Frequency Coupled Power BRIDGE 1 BRIDGE 1 f BROADBAND RESULTANT Decoupled Power frequency Coupled Foupled j ~~~Power ~Power Low Pass Decoupled | Filter |coupled Power Frequency-. Frequency BRIDGE 2 FIG. 2-27: BROADBANDING BY 2 BRIDGE LINKS WITH FILTERS 39

THE UNIVERSITY OF MICHIGAN 7692-3 —Q 2.6 Coupling Between a Spiral and a Slot A ground plane has been made for studying coupling and decoupling between S-band antennas such as slots, spirals and monopoles. Special attention was given to making some holes in the ground plane as closely together as possible so that the near field coupling effects can be included. As a preliminary work to future decoupling study, the coupling data between a spiral and a slot were obtained. The spirals used were two AFA circular Archimedean spirals and two Advanced Development Laboratories square Archimedean spirals. The arrangement of the antenna on the ground plane is shown in Fig. 2-28. It is obvious from this figure that there are four factors affecting the value of coupling. These are: 1) The relative orientation of the slot with respect to a fixed spiral. 2) The center-to-center distance between the two antennas. 3) The azimuth angle 0 or orientation of the spiral. 4) The frequency. First consider the coupling along the two axes A-A and B-B which will be designated as the E -plane and H-plane coupling respectively. This should not be considered as inadequate to demonstrate the nature of the coupling since the coupli on the two axes are respectively the maximum and the minimum values and therefore yield the most informative data. Figures 2-29 to 2-32 show the results of circular spiral to slot coupling. Although the coupling was plotted as a function of 0 with f and d as parameters, its dependence on f and d can readily be seen by comparing the data on different curves. The effect of spiral orientation relative to the slot can be summarized. The E-plane coupling is seen to be stronger than H-plane coupling, especially for larger distances owing to the fact that the near field term is largely responsible for Hplane coupling. 40

THE UNIVERSITY OF MICHIGAN 7692-3-Q m Ct~~~~~~~~~~~~~~~~C I 0 r~~~~~~~ m~~~ u~~~~~~~~~~~~~~~~~~ ^ O _ _ _ z m~~~~ 41.

-20 ~ 2. 2 GHz O | 2.6 GHz -40 |x 3. 0 GHz ~-40 o 3.8 GHz.. I I i I 0 90 180 270 360 (Degree) FIG. 2-29: E-PLANE COUPLING BETWEEN A CIRCULAR SPIRAL AND A SLOT VERSUS AZIMUTH ANGLE. (Center-to-center Distance 8.5 cm) i~~~~~~~~~~~~~~~~~~~~~~~5

-G~~~~~~~~~z ~~~~~~~~ -30 rNr (0' CD~~~~~~ 0 2.2 GHz 0o 2.6 GHz x 3.0 GHz A 3.4 GHz 0 3.8 GHz -50 0 0 90 180 270 360 0 (Degree) FIG. 2-30: E-PLANE COUPLING BETWEEN A CIRCULAR SPIRAL AND A SLOT VERSUS AZIMUTH ANGLE. (Center-to-center Distance 26 cm).

-30 F P-40 2.2 GHz 0 2.6 GHz X 3.0 GHz ) 3.4GHz o 3.8GHz -50 __ _... 0 90 180 270 360 (Degree) FIG. 2-31: H-PLANE COUPLING BETWEEN A CIRCULAR SPIRAL AND A SLOT VERSUS AZIMUTH ANGLE. (Center-to-center Distance 9.6 cm).

-40 -50 r~~~~-f 3 I I CI o ) 0 -60 2.2 GHz o 2.6 GHz X 3.0 GHz A 3.4 GHz o 3.8 GHz 0 0 90 180 270 360 (Degree) FIG. 2-32: H-PLANE COUPLING BETWEEN A CIRCULAR SPIRAL AND A SLOT VERSUS AZIMUTH ANGLE. (Center-to-center Distance 25.5 cm).

THE UNIVERSITY OF MICHIGAN 7692-3-Q The 0 variation of coupling is largely dependent on frequency. For 2. 2 GHz and 2. 6 GHz the variation is small; for 3. 4 GHz and 3. 8 GHz it is very large; for 3. 0 GHz it is intermediate. This indicates clearly the influence of the less symmetrical structure of the spiral on coupling because the radiation region approaches the center of the spiral as the frequency goes up to the high frequency limit of the spiral. For 2. 2 GHz and 2. 8 GHz the radiation region has a radius about one-half that of the spiral. Good symmetry and small reflections from the end result in almost omnidirectional coupling. The center-to-center distance and frequency dependence can be simply stated. These are intimately related. It is hard to draw any conclusion before normalizing to coupling versus wavelengths spacing. However, it is seen that the coupling does not follow the far field 6 db per octave rule for both E-plane and H-plane coupling. Most of the features in the spiral-to-slot coupling are not unlike those previously published by this project group on spiral-to-monopole coupling. For an interesting comparison, the reader is referred to earlier reports (J.A. M. Lyon, et al, 1966). 46

THE UNIVERSITY OF MICHIGAN 7692-3-Q III ABSORBING MATERIALS In this report magnetic loss tangent and magnetic Q data are included completing a twelve sample series (Fig. 3-1 through 3-5). The frequency range where measurements were taken necessitated the determination of the I' and p" parameters by the use of an impedance bridge and a sample holder shorted at one end with the specimen in contact with the short termination. Ideally if the sample is placed in the position where the electric field is maximum, or at X/4 away from the short termination one should be able to determine c' and e". However, due to the fact that one is unable to obtain with this technique reproducible results at the particular frequency range of 50 - 400 MHz the data obtained on E' and e" are not reported. Concurrently with the above effort, measurements are taken now at frequencies between 2000 - 7000 MHz. The technique used for this particular range of frequencies is that of the determination of the propagation constant y and intrinsic impedance of the specimen through the determination of open and short circuit impedance measurements with the sample in a coaxial line. This method appears to be the most accurate one used so far since corrections compensate for the residual VSWR on the line and the noise level. A computer program is available for the calculations. From the information presented in this report one may deduce that some of the mixtures present a high absorbing capacity. Of particular notice is mixture 2 which has both broadband characteristics and high magnetic loss tangent between 50 and 400 MHz as can be seen in Fig. 3-2b. A complete chart of the data appears in Table IV-1. 47

Q Q/m2 20 4 15 / / \ 1 2 zd0 10 10 ~ / C 2 (a) (b) 5 - I I' I I I I I ' i a ' ' ' ' 'a' ' b Z 100 150 200 250 300 350 400 100 150 200 250 300 350 400 f (MHz) f (MHz) (c) (d) FIG. 3-1: SPECIMEN MAGNETIC Q VERSUS FREQUENCY. 0~~~~~~~~~~~ 0\ \0~~~~~~~~ FIG. 3-1: SPECIMEN MAGNETIC Q VERSUS FREQUENCY.

tan 6 m 1) tan 6 S H 0.3 0.75 / 0.50 0.2 \ 0.25 0. 1 w I I I I - I I I I0I.251 100 150 200 250 300 350 400 100 150 200 250 300 350 400 f (MHz) f (MHz) tans6 (a) tan 6 (b) m3 mA 1.0 0I * 1.5 0.5 1.0 00 C 0. 5 * 0 00.* 0 I I 1 1 I I~~~~s a I I I L LI I 100 150 200 250 300 350 400 100 150 200 250 300 350 400 f (MHz) f (MHz) (c) (d) FIG. 3-2: SPECIMAN LOSS TANGENT VERSUS FREQUENCY

9m5 Em strQ 5 6 2.5 2.0 2 rl 1.5 L33SE M MC~ V N* * H 1.0 1 ' '/ — ' ' 0.5 100 150 200 250 300 350 400 100 150 200 250 300 350 400 f (MHz) f (MHz) (a) (b) 0 FIG. 3-3: SPECIMEN MAGNETIC Q-FACTOR VERSUS FREQUENCY.

tan 6 tan 6 m 6 5 H 1.2 1. 0 0.8 I 1.0 - 0.6 -0 1 U0.4 - 0.5 03 0. 2o 100 150 200 250 300 350 400 100 150 200 250 300 350 400 f (MHz) f (MHz) (a) (b) FIG. 3-4: SPECIMEN MAGNETIC LOSS TANGENT VERSUS FREQUENCY. Z

tan 6 tan 6 n}2 m 12 150 100 / )0.1 * x1 01 ~I,,, | I I I A I I I I I ' I 100 150 200 250 300 350 400 100 150 200 250 300 350 400 f (MHz) f (MHz) ) (a) (b) FIG. 3-5: (a) MAGNETIC Q VERSUS FREQUENCY, (b) MAGNETIC LOSS TANGENT VERSUS Z FREQUENCY.

TABLE Il-1 H f C~ I tan5 ',tan6 Q tan6 Q tan6 IQ tan6 Q tan6 fm Q'I ta m ma m6 m i tan6b (MHz) i 1 4 m m5 m2 m12 m m~~~~~~~~~~~~ 6 m2 12 75 4.5 0. 222 2.74 0. 365 1.43 0,7 1.57 0. 636 2.15 0. 465 1.43 0. 7 33.6 0.0298 Z 100 11.4 0.097 3.27 0. 306 1, 73 0.578 1,25 0.8 2.15 0.465 1.07 0.935 22. 9 0.0437 125 19.2 0.052 2.6 0.385 1.43 0.7 1.19 0.84 1.73 0.6 1.425 0.7 28.6 0.035 ~J - Q1a 150 3.49 0.286 1.3 0.77 1.92 0.522 1.0 1.0 1.07 0.935 1.2 0.834 9.5 0.105 H 200 7.62 0.131 1.28 0.775 2.14 0.468 1.1 0.91 1.3 0.768 1.0 1.0 28.6 0.035 250 3.72 0. 269 1.29 0. 753 1,28 0,78 1.92 0.52 1.425 0.7 1.735 0.577 28.75 0.0346 300 7.65 0.130 1.3 0.77 1.33 0.752 2.14 0.467 1.19 0.84 1.0 1.0 57.2 0.0174 350 3.74 0. 268 1.3 0,77 1,43 0.7 2.14 0.467 0. 921 1.087 2.75 0. 363 143 0.007 400 3.18 0.315 1.3 0.77 0.87 1.145 0.919 1.08 1.19 0.84 0.912 1.095 19.15 0.056 __ __ __ __ _ __ __ __ _ __ __ __ __ _ __ __ __ _ __ __ __ _

THE UNIVERSITY OF MICHIGAN 7692-3-Q IV CONCLUSIONS The work effort during this report period has indicated promising improvement of isolation obtained over what was reported for the previous quarter. The 15 db of isolation acquired through the use of abridge link, goodfor 1. 5 GHz bandwidth coverage for the X-band, is indeed promising. However, the use of absorbing material with E-Sectoral Horns providing 18 db of isolation system to system over X-band is also encouraging. Circumferential chokes around a driven slot or an active slot giving 22 db system-to-system over 0. 5 GHz can offer a useful isolation increase if this narrow bandwidth coverage is satisfactory for X-band. The same method offers 14 db increased isolation over the entire X-band. The work reported on corrugations is still preliminary. Much needs to be done to make corrugations provide a satisfactory increase of isolation. Likewise, the use of parasitic slot elements has brought on a system-to-system basis an increase of 7 db over 2.5 GHz for X-band. The decoupling results are summarized in Table IV-1 of this section. This table enables a simple comparison to be made of the various methods. In actual use of methods to combat a known system-to-system interference problem, it is quite likely that more than one of the several methods will be needed to bring the offending signal down to an appropriate level. It is to be observed in this table that proper initial orientation of antennas within the limitations of polarization and direction requirements of the antennas affords a very useful means of introducing isolation between two systems. The other methods in the table are by their nature suited to modifications. However, the initial antenna orientation is one dictated by the design requirements for operation of the systems involved. All coupling reduction techniques are adaptable to systems operating at various frequency bands. 54

Isolation Method Antennas Used System Isolation Bandwidth Side Effects Increase in db+ Antenna Slots, E- and H- At least 40 X-band None Orientation Sectoral Horns RF Bridge Slots 15 1.5 GHz Up to 5 db isolation decrease for higher and lower frequencies. Antennas Recessed in E - Sectoral Horn 18 db X-band Up to 0. 6 db loss in maximum Z Cavity Partially Filled Slots 13 db X-band Up to 1 db increase in maximum gain 'With Absorber Half-power beamwidth reduced. Circumferential 22 db 0. 5 GHz Up to 4 db increase in maximum gain Chokes Slots14 db X-band Half-power beamwidth reduced. - Corrugations Slots 5 db X-band Assymmetrical radiation pattern with 2 db troughs within ~ 30 of broad-0 side. Parasitic Antenna Slots 7 db 2. 5 GHz Assymmetrical radiation pattern. C) TABLE IV-1: SUMMARY OF DECOUPLING METHODS AND RESULTS. (ALL STRUCTURES ARE FLUSH-MOUNTED.) +For E-plane coupling Z C+All experiments conducted in X-band (8. 2 - 12.4 GHz) Theoretically infinite. i(Gain) = (Directivity) x (Efficiency)

THE UNIVERSITY OF MICHIGAN 7692-3-Q The various methods of increasing decoupling or increasing isolation that have been studied in this report have been made with full cognizance of the undesirability of modifying the radiation characteristics of the antennas involved. For this reason, experimental data have been shown indicating to some extent the limitations impose on the radiation pattern of an antenna by the influence of absorber or surface modifications.

THE UNIVERSITY OF MICHIGAN 7692-3-Q v FUTURE EFFORT During the next report period, it is anticipated that extensive work will be done on the bridge link method which has been described in this report. It is expected that additional bridge links will be utilized each, one covering a given bandwidth. By the use of several links, each with its own filtering, it is expected that the entire X-band could be covered by this method. The methods using absorber will benefit by the purchase of different types of absorber and by the use of absorber which has been fabricated on this project. The fabrication of absorber on this project has been limited to readily mixed materials having the desired electrical characteristics. It is expected that isolation by either chokes or corrugations will be improved both in magnitude and bandwidth by the use of appropriate absorbing material. ACKNOWLE DGME NTS Mr. D.R. Brundage of The University of Michigan, Institute of Science and Technology, obtained most of the experimental data in this report. 57

THE UNIVERSITY OF MICHIGAN 7692-3-Q REFERENCES Coe, R. J. and Held, G. (January, 1964), "A Parasitic Slot Array, " IEEE Trans., AP-12, No. 1, pp. 10-16. Hurd, R.A. (December, 1954), "The Propagation of an Electromagnetic Wave Along an Infinite Corrugated Surface, " Can. J. Phys., 32, No. 12, pp. 727-734. Kraus, J. D. (1950), Antennas, McGraw-Hill Book Company. Lyon, J.A. M., Kalafus, R. M., et al (April, 1966), "Derivation of Aerospace Antenna Coupling-Factor Interference Prediction Techniques, " The University of Michigan Radiation Laboratory, Technical Report AFAL-TR-66-57. Lyon, J.A. M., et al (May, 1966), "Electromagnetic Coupling Reduction Techniques"' The University of Michigan Radiation Laboratory, Report 7692-2-Q. 58

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THE UNIVERSITY OF MICHIGAN DISTRIBUTION LIST (continued) ASD (AS ZIE) Wright-Patterson AFB Ohio 45433 ASD (ASZW) Wright-Patterson AFB Ohio 45433 ASD (ASTFP) Wright-Patterson AFB Ohio 45433 SEG (SEACR) Wright-Patterson AFB Ohio 45433 AFIT Wright-Patterson AFB Ohio 45433 Electromagnetic Compatibility Analysis Center Code ACG U.S. Navy Marine Engineering Laboratory Anapolis, Maryland 21402 Electromagnetic Compatibility Analysis Center Code ACZ U.S. Navy Marine Engineering Laboratory Anapolis, Maryland 21402 Electromagnetic Compatibility Analysis Center Code ACO U.S. Navy Marine Engineering Laboratory Anapolis, Maryland 21402 RADC (EMATA) Griffiss AFB New York 13442 1 (continued)

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THE UNIVERSITY OF MICHIGAN DISTRIBUTION LIST (continued) Ohio State University Research Foundation Attn: Technical Library (M/F Antenna Lab) 2024 Neil Avenue Columbus, Ohio 43210 1 Polytechnical Institute of Brooklyn Attn: Professor A. A. Oliner Microwave Research Institute 55 Johnson Street Brooklyn, New York 11201 1 Stanford Electronic Laboratory Attn: Librarian (Antenna Laboratory) Stanford, California 94025 1 Syracuse University Electrical Engineering Department Attn: Dr. Jose Perihi Syracuse, New York 13210 1 University of Southern California EE Department, Mr. W. V. Trusch University Park Los Angeles, California 90007 1 The University of Michigan Radiation Laboratory, Dept. of EE. R. E. Hiatt 201 Catherine Street Ann Arbor, Michigan 48108 1 Total 100 copies

UNCLASSIFIED Security Classification DOCUMENT CONTROL DATA- R&D (Security classificatton of title, body of abstract and indexing annotation must be entered when the overall report is classified) 1. ORIGINATIN G ACTIVITY (Corporate author) aa. REPORT SECURITY C LASSIFICATION The University of Michigan Radiation Laboratory Unclassified Department of Electrical Engineering 2 b ROJP 3. REPORT TITLE Electromagnetic Coupling Reduction Techniques 4. DESCRIPTIVE NOTES (Type of report and inclusive date.) Third Quarterly Report - 15 June 1966 - 14 August 1966 5. AUTHOR(S) (Lset name, first name, inttjal) Lyon, John A. M., Alexopoulos, Nicholas G., Cha, Alan G., Digenis, Constantine J., and Parker, William W. 6. REPORT ATE,. 7a. TOTAL NO. OF PAGES 7b. NO. OF REFS August 1966 58 5 8a. CONTRACT OR GRANT NO. I4. ORIG~INATOR'8 REPORT NUMBER(S) AF 33(615)-3371 b. pROJECT NO 7692-3-Q 4357 c. 9b. OTH ER REPORT NQ('S) (Any other numbers that may be atsigned this report) 435709 d. 10. A'v A IL ABILITY/!MITATION, NOTICES Qualified requestors may obtain copies of this report from DDC. This document is subject to special export controls and each transmittal to foreign governments or foreign nationals may be made only with prior approval of AFAL(AVPT, WrightPatterson AFB, Ohio. 11. SUPPI. lEMENTARY NOTES 11..PQNSORING M[l.ITARY ACTIVITY Air Force Avionics Laboratory AVWE Research and Technology Division, AFSC Wright-Patterson AFB, Ohio 45433 13. ABSTRACT During this report period, additional studies were made on horns and slots surrounded by absorbing material. New designs were used which permitted all of the flush mounting material to be flush with the conducting surface. Swept frequency methods were used in obtaining values for the increased isolation over a considerable range of frequency. Studies were made on the isolation of two antennas through the use of chokes. A slot antenna was constructed surrounded by four chokes in the form of circumferential trenches. For two systems, each equipped with such circumferential chokes, an additional 22 db of isolation was obtained. Additional studies were made on the use of corrugations to increase isolations. The corrugations described here are not of an optimized design and do not use absorber material as has been planned for the future. The use of simple parasitic antenna elements, such as slots as reflectors, has been used to improve system isolation. The improvement is rather small, being limited to about 7 db. Considerable effort has been utilized on developing an RF bridge to improve isolation by cancellation of the unwanted signal. Results have been encouraging but work remains to adapt this method to a wider bandwidth. DD. JAN 64 1473 UNCLASSIFIED Security Classification

UNCLASSIFIED Security Classification 14. LINK A LINK B LINK C KEY WORDS ROLE WT ROLE WT ROLE! WT Coupling Decoupling Isolation increase INSTRUCTIONS 1. ORIGINATING ACTIVITY: Enter the name and address imposed by security classification, using standard statements of the contractor, subcontractor, grantee, Department of De- such as: fense activity or other organization (catporate author) issuing (1) "Qualified requesters may obtain copies of this (1) "Qualified requesters may obtain copies of this ~~~~~~~~~~the report. ~report from DDC." 2a. REPORT SECURITY CLASSIFICATION: Enter the over2a. REPORT SECU.TY CLASSIFICATION: Enter the over- (2) "Foreign announcement and dissemination of this all security classification of the report. Indicate whether "Restricted Data" is included. Marking is to be in accordance with appropriate security regulations. (3) "U. S. Government agencies may obtain copies of this report directly from DDC. Other qualified DDC 2b. GROUP: Automatic downgrading is specified in DoD Di- users shall request through rective 5200. 10 and Armed Forces Industrial Manual. Enter the group number. Also, when applicable, show that optional markings have been used for Group 3 and Group 4 as author- (4) "U. S. military agencies may obtain copies of this report directly from IDC. Other qualified users 3. REPORT TITLE: Enter the complete report title in all shall request through capital letters. Titles in all cases should be unclassified. If a meaningful title cannot be selected without classificu- ' tion, show title classification in all capitals in parenthesis (5) "All distribution of this report is controlled, Qualimmediately following the title. ified DDC users shall request through 4. DESCRIPTIVE NOTES: If appropriate, enter the type of, report, e.g., interim, progress, summary, annual, or final. If the report has been furnished to the Office of Technical Give the inclusive dates when a specific reporting period is Services, Department of Commerce, for sale to the public, indicovered. cate this fact and enter the price, if known. 5. AUTHOR(S): Enter the name(s) of author(s) as shown on 11. SUPPLEMENTARY NOTES: Use for additional explanaor in the report. Enter last name, first name, middle initial. tory notes If military, show rank and branch of service. The name of the principal au.thor is an absolute minimum requirement. 12. SPONSORING MILITARY ACTIVITY: Enter the name of the departmental project office or laboratory sponsoring (pay-;6. REPORT DATE. m Enter the date of the report as day, ing for) the research and development. Include address. month, year; or month, year. If more than one date appears on the report, use date of publication. 13. ABSTRACT: Enter an abstract giving a brief and factual summary of the document indicative of the report, even though 7a. TOTAL NUMBER OF PAGES: The total page count it may also appear elsewhere in the body of the technical reshould follow normal pagination procedures, i.e., enter the port. If additional space is required, a continuation sheet shall number of pages containing information. be attached. 7b. NUMBER OF REFERENCES: Enter the total number of It is highly desirable that the abstract of classified reports references cited in the report. be unclassified. Each paragraph of the abstract shall end with 8a. CONTRACT OR GRANT NUMBER: If appropriate, enter an indication of the military security classification of the inthe applicable number of the contract or grant under which formation in the paragraph, represented as (TS), (S), (C), or (U). the report was written. There is no limitation on the length of the abstract. How8b, 8c, & 8d. PROJECT NUMBER: Enter the appropriate ever, the suggested length is from 150 to 225 words. military department identification, such as project number, subproject number system numbers task number et14. KEY WORDS: Key words are technically meaningful terms subproject number, system numbers, task number, etc. or short phrases that characterize a report and may be used as 9a. ORIGINATOR'S REPORT NUMBER(S): Enter the offi- index entries for cataloging the report. Key words must be cial report number by which the document will be identified selected so that no security classification is required. Identiand controlled by the originating activity. This number must fiers, such as equipment model designation, trade name, military be unique to this report. project code name, geographic location, may be used as key 9b. OTHER REPORT NUMBER(S): If the report has been words but will be followed by an indication of technical conassigned any other report Eumbers (either by the originator text. The assignment of links, rules, and weights is optional. or by the sponsor), also enter this number(s). 10. AVAILABILITY/LIMITATION NOTICES: Enter any lim- I itations on further dissemination of the report, other than those UNCLASSIFIED Security Classification

UNIVERSITY OF MICHIGAN 111151 03465 868511111111111 3 9015 03465 8685